Photonic serial digital-to-analog converter employing a heterojunction thyristor device

ABSTRACT

A serial photonic digital-to-analog converter employs a heterojunction thyristor device configured for optically-controlled sampling/switching to convert a digital word encoded by a serial digital optical data signal (e.g., serial optical bit stream) into a corresponding analog electrical signal. A voltage reference is operably coupled to the electrical input terminal of the heterojunction thyristor device. The voltage reference cooperates with the heterojunction thyristor device to sequentially generate at its electrical output terminal a voltage signal representing contribution of each bit of the digital word encoded in the serial digital optical data signal. A summing network is operably coupled to the electrical output terminal of the device. The summing network sequentially sums contribution of the voltage signal over the sequence of bits to produce an analog electrical signal corresponding to the digital word for output therefrom. Preferably, the summing network includes an adding node, sample and hold circuit, and a feedback path between the sample and hold circuit and the adding node. In addition, the voltage reference preferably supplies a voltage level corresponding to the maximum voltage level of the analog electrical signal divided by 2 (N−1) , where N is the number of bits in said digital word, and the feedback path comprises an amplifier that amplifies output of the sample and hold circuit by a factor of 2.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a continuation-in-part of U.S. application Ser. No.10/280,892, filed Oct. 25, 2002, entitled “Optoelectronic DeviceEmploying At Least One Semiconductor Heterojunction Thyristor ForProducing Variable Electrical/Optical Delay,” commonly assigned toassignee of the present invention, and herein incorporated by referencein its entirety.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates broadly to the field of optoelectronics devices,and, more particularly to optoelectronic circuits that convert opticalsignals into electrical signals and/or perform high speed signalsampling operations.

2. State of the Art

Optical networks provide the advantages of increased speed andtransmission capacity for carrying voice and data. In optical networks,optical signals (e.g., light waves) are used to carry the informationover the network. This information is provided by a source typically inelectrical form and converted into an optical signal for transmissionover the network.

In order to carry voice, the sound waves of the spoken voice aretypically converted into an analog electrical signal which is convertedinto a digital electrical signal consisting of bits of information,wherein each bit is either a logic level ‘1’ (the amplitude of thedigital electrical signal is high/ON) or logic level ‘0’ (the amplitudeof the digital electrical signal is low/OFF). Data is typically storedin digital form as bits of information, and thus analog-to-digitalconversion is not necessary.

Once the digital electrical signal has been obtained, it is converted toa digital optical signal by an electrical-to-optical converter, whichmodulates a laser light source in response to the digital electricalsignal. The digital optical signal consists of bits of information,wherein each bit is either a logic level ‘1’ (the light intensity levelof the digital optical signal is high/ON) or a logic level ‘0’ (thelight intensity level of the digital optical signal is low/OFF). Thedigital optical signal is then transmitted over a medium (such as alight guide optical fiber).

An optical-to-electrical converter receives the digital optical signalproduced by the laser source and transmitted over the medium, andgenerates a digital electrical signal corresponding to the digitaloptical signal. The digital electrical signal may be converted back to adigital optical signal for transmission over the optical network (suchas the case where the optical-to-electrical converter is part ofswitch/router that operates in the electric domain on digital electricalsignals). Alternatively, the digital electrical signal may be convertedto an analog electrical signal (such as the case for voice applicationswhere the analog electrical signal is converted back into sound wavesthat can be interpreted by the person receiving the phone call).Moreover, the digital electrical signal may be transformed forcommunication over a data communication link (such as the case for dataapplications where the digital electrical signal is transformed (e.g.packetized) for communication over a data communication link, such as aGigabit Ethernet link).

FIG. 1A is a prior art functional block diagram illustrating a typicaloptical-to-electrical converter including a photodetector 110 (which maybe one or more avalanche photodiodes or one or more PIN photodiodes)that converts the light level of the received digital optical signal toa signal current. The photodetector 110 delivers the extracted currentto a transimpedance amplifier (TIA) 111, which first converts thecurrent to a voltage. This single-ended voltage is amplified by the TIAand typically converted to a differential signal. A post amplifier 112is provided, which in most cases is configured as a limiting amplifierthat delivers a certain output-voltage swing whose maximum isindependent of the input signal strength. A data recovery circuit 113performs amplitude-level analysis on the signal output by post amplifier112 to recover the serial digital data signal (in electrical form) fromthe received optical signal. Demultiplexing circuit 114 performs aserial-to-parallel conversion on the serial digital data streamgenerated by the data recovery circuit 113 to generate a multi-bitdigital signal (electrical) representing a sequence of bits in thereceived digital optical signal.

The mechanism of FIG. 1A that converts the digital optical signal to adigital electrical signal (the photodetector 110, TIA 111, postamplifier 112 and data recovery circuit 113) is costly to design andmanufacture because of the complex nature of the TIA 111, post amplifier112 and data recovery circuit 113, and because of difficulties inintegrating one or more of these components with the photodetector 110.

Thus, there is a great need in the art for an optoelectronic circuitthat converts a digital optical signal to a digital electrical signal ina manner that has lower cost and improved ease of integration, and thatis suitable for high speed applications.

The limitations of FIG. 1A are also present in parallel optical datalinks that have been developed to provide for increased aggregate datarates. As shown in prior art FIG. 1B, a parallel optical data linkconsists of a transmit module 120 coupled to a receive module 122 with amulti-fiber connector 124. The transmit module typically employs anarray 126 of vertical-cavity-surface-emitting lasers (VCSELs) and amulti-channel laser driver integrated circuit 128 for driving the arrayof lasers to produced a plurality of synchronous optical bit streamsthat are transmitted over the multi-fiber connector 124. The receivemodule 122 includes a photodetector array 130 (typically realized withP-I-N diodes) that receives the synchronous optical bit streams andcooperate with an integrated circuit 132 that provides a correspondingarray of low noise transimpedance amplifiers, limiting amplifiers, anddata recovery circuits to produce a plurality of electrical bit streamscorresponding thereto. The plurality of electrical bit streams areprovided to one or more integrated circuits 134 that map parallel bitsencoded in the plurality of electrical bit streams into a predetermineddata format (such as a SONET frame). Although such a parallel opticaldata link provides cost savings based upon array-integration ofelectronic and optoelectronic components in both the transmit module andthe receive module, it suffers from the same limitations of the approachof FIG. 1A. The complex nature of the TIA, post amplifier and datarecovery circuit in the receive module 122 leads to increased designcosts and manufacture costs of the receive module 122, and also leads todifficulties in integrating one or more of these components with thephotodetector array as part of the receive module 122.

Thus, there is a great need in the art for an optoelectronic circuitthat converts a plurality of synchronous optical bit streams toelectrical bit streams in a manner that has lower cost and improved easeof integration, and that is suitable for use in high speed applications(such as the receive module of a parallel optical data link).

For high frequency applications, optoelectronic integrated circuits thatconvert an optical signal to an electric signal have been reported. Forexample, Dutta et al., “10 GHz bandwidth monolithic p-i-nmodulation-doped field effect transistor photoreceiver,” Appl. Phys.Lett., Vol. 63, No. 15, Oct. 1993, pp. 2115-2116, describes the use anInGaAs PIN photodiode for the conversion of incident photons toelectrons followed by an amplifier circuit based on a modulation-dopedfield effect transistor. And Akahori et al., “10-GB/s High-SpeedMonolithically Integrated Photoreceiver Using InGaAs p-i-n PD and PlanarDoped InAlAs/InGaAs HEMT's,” IEEE Photonics Technology Letters, Vol. 4,No. 7, Jul. 1992, pp. 754-756 describes the use of an InGaAs PINphotodiode for the conversion of incident photons to electrons followedby an amplifier circuit based on planar doped InAlAs/InGaAs HEMTdevices. And Hurm. et. al., “20 Gbit/s long wavelength monolithicintegrated photoreceiver grown on GaAs,” Electronics Letters, Vol. 33,No. 7, 1997, pp. 624-626, describes the use an MSM photodiode for theconversion of incident photons to electrons followed by an amplifiercircuit based on an AlGaAs/GaAs HEMT transistor. However, these priorart mechanisms require substantially different epitaxial growthstructures to realize the components of the optoelectronic integratedcircuit, and thus are costly to design and manufacture.

As described above, digital optical signals may be used to carry analoginformation (such as voice). In such applications, it is necessary thatthe optical bits encoded in the digital optical signal be converted intoan analog electrical signal for subsequent processing. Prior art FIG. 1Cis a functional block diagram illustrating a typical mechanism forperforming such conversion operations. Similar to FIG. 1A, aphotodetector 110 converts the light level of the received digitaloptical signal to a signal current. The photodetector 10 delivers theextracted current to a transimpedance amplifier (TIA) 111, which firstconverts the current to a voltage. This single-ended voltage isamplified by the TIA and typically converted to a differential signal. Apost amplifier 112 is provided, which in most cases is configured as alimiting amplifier that delivers a certain output-voltage swing whosemaximum is independent of the input signal strength. A data recoverycircuit 113 performs amplitude-level analysis on the signal output bypost amplifier 112 to recover the serial digital data signal (inelectrical form) from the received optical signal. Demultiplexingcircuit 114 performs a serial-to-parallel conversion on the serialdigital data stream generated by the data recovery circuit 113 togenerate a multi-bit digital signal (electrical) representing a sequenceof bits in the received digital optical signal. The multi-bit digitalsignal produced by the demultiplexing circuit 114 is provided to adigital-to-analog converter 115 that converts the multi-bit digitalsignal to a corresponding analog electrical signal. This approachsuffers from the same limitations of the approach of FIG. 1A, whereinthe complex nature of the TIA, post amplifier and data recovery circuitleads to increased design costs and manufacture costs. In addition, thelarge number of complex components that make up the signal processingchain (from photodetector 110 to the digital-to-analog converter 115)are costly to design and manufacture.

Thus, there is a great need in the art for an optoelectronic circuitthat converts a digital optical signal to an analog electrical signal ina manner that has lower cost and improved ease of integration, and thatis suitable for high speed applications.

Similarly, a parallel optical data link may be used to carry analoginformation (such as voice). In such applications, it is necessary thatthe optical bits encoded in the digital optical signals be convertedinto an analog electrical signal for subsequent processing. Thisapproach suffers from the same limitations of the approach of FIGS. 1Aand 1B, wherein the complex nature of the TIA, post amplifier and datarecovery circuit leads to increased design costs and manufacture costs.In addition, the large number of complex components that make up thesignal processing chain (from photodetector to the digital-to-analogconverter) are costly to design and manufacture.

Thus, there is a great need in the art for an optoelectronic circuitthat converts parallel optical bit streams to an analog electricalsignal in a manner that has lower cost and improved ease of integrationand that is suitable for high speed applications.

In addition, digital-to-analog converters (and other signal processingcircuitry such as analog-to-digital converters, switched-capacitancefilters/amplifiers, and switched-capacitance waveform generators)typically employ electrically-controlled transistors as on-off switchesto perform signal sampling operations. Due to parasitic capacitance andintrinsic capacitances between the input and output nodes of thesampling transistor, feedthrough charge that collects on the sampledsignal increases to an intolerable level at high frequencies. Therefore,the electronic sampling technique becomes limited in sensitivity at highsampling rates.

Thus, there is a great need in the art for improved signal samplingmechanisms that are suitable for high sampling rates and avoid thelimitations (including feedthrough charge) of the prior arttransistor-based sampling mechanisms.

SUMMARY OF THE INVENTION

It is therefore an object of the invention to provide an improvedoptoelectronic circuit that converts a digital optical signal to adigital electrical signal in a manner that avoids the limitations of theprior art.

It is another object of the invention to provide an improvedoptoelectronic circuit that converts a digital optical signal to adigital electrical signal in a manner that has lower cost and improvedease of integration.

It is another object of the invention to provide an improvedoptoelectronic circuit that converts a digital optical signal to adigital electrical signal in a manner that is suitable for high speedapplications.

It is another object of the invention to provide an improvedoptoelectronic circuit that converts a digital optical signal to adigital electrical signal in a manner that affords ease of integrationwith a broad range of devices (such as FET transistors, bipolartransistors, lasers, optical modulators, and waveguide devices).

It is a further object of the invention to provide an improvedoptoelectronic circuit that utilizes a heterojunction thyristor devicethat is adapted to receive a digital optical signal and convert thedigital optical signal to a digital electrical signal.

It is an additional object of the invention to provide a plurality ofsuch optoelectronic circuits to convert a plurality of synchronousoptical bit streams to electrical bit streams in a manner that issuitable for high speed applications, such as in the receive module of aparallel optical data link.

It is an additional object of the invention to provide an improvedoptoelectronic circuit that converts a parallel digital optical signal(that synchronously encodes a plurality of bits of information) to ananalog electrical signal in a manner that avoids the limitations of theprior art.

It is another object of the invention to provide an improvedoptoelectronic circuit that converts a parallel digital optical signalto an analog electrical signal in a manner that has lower cost andimproved ease of integration.

It is another object of the invention to provide an improvedoptoelectronic circuit that converts a parallel digital optical signalto an analog electrical signal in a manner that is suitable for highspeed applications.

It is another object of the invention to provide an improvedoptoelectronic circuit that converts a parallel digital optical signalto an analog electrical signal in a manner that affords ease ofintegration with a broad range of devices (such as FET transistors,bipolar transistors, lasers, optical modulators, and waveguide devices).

It is a further object of the invention to provide an improvedoptoelectronic circuit that utilizes a plurality of heterojunctionthyristor devices to convert a parallel digital optical signal to ananalog electrical signal.

It is an additional object of the invention to provide an improvedoptoelectronic circuit that converts a serial digital optical signal(that serially encodes a plurality of bits of information) to an analogelectrical signal in a manner that avoids the limitations of the priorart.

It is another object of the invention to provide an improvedoptoelectronic circuit that converts a serial digital optical signal toan analog electrical signal in a manner that has lower cost and improvedease of integration.

It is another object of the invention to provide an improvedoptoelectronic circuit that converts a serial digital optical signal toan analog electrical signal in a manner that is suitable for high speedapplications.

It is another object of the invention to provide an improvedoptoelectronic circuit that converts a serial digital optical signal toan analog electrical signal in a manner that affords ease of integrationwith a broad range of devices (such as FET transistors, bipolartransistors, lasers, optical modulators, and waveguide devices).

It is a further object of the invention to provide an improvedoptoelectronic circuit that utilizes at least one heterojunctionthyristor device to convert a serial digital optical signal to an analogelectrical signal.

It is an additional object of the invention to provide an improvedoptoelectronic circuit that provides optically-controlled (orelectrically-controlled) signal sampling in a manner that avoids thelimitations of the prior art.

It is another object of the invention to provide an improvedoptoelectronic circuit that provides optically-controlled (orelectrically-controlled) signal sampling in a manner that has lower costand improved ease of integration.

It is another object of the invention to provide an improvedoptoelectronic circuit that provides optically-controlled (orelectrically-controlled) signal sampling in a manner that is suitablefor high speed applications.

It is another object of the invention to provide an improvedoptoelectronic circuit that provides optically-controlled (orelectrically-controlled) signal sampling in a manner that affords easeof integration with a broad range of devices (such as FET transistors,bipolar transistors, lasers, optical modulators, and waveguide devices).

It is a further object of the invention to provide an improvedoptoelectronic circuit that utilizes a heterojunction thyristor deviceto perform optically-controlled (or electrically-controlled) signalsampling.

In accordance with the objects of present invention, a heterojunctionthyristor device is configured to convert an input digital opticalsignal to an output digital electrical signal. The input digital opticalsignal encodes bits of information (each bit representing an OFF logiclevel or ON logic level) and is part of the Optical IN signal that isresonantly absorbed by the device. A sampling clock defines samplingperiods that overlap the bits (e.g., ON/OFF pulse durations) in theinput digital optical signal. The sampling clock can be in the form ofelectrical pulses supplied to the n-channel injector terminal(s) and/orp-channel injector terminals of the heterojunction thyristor device.Alternatively, the sampling clock can be in the form of optical pulsesthat are part of the Optical IN signal that is resonantly absorbed bythe device. The heterojunction thyristor device operates in an OFF stateand an ON state. In the OFF state, current does not flow between ananode terminal and a cathode terminal of the device; while in the ONstate, current flows between the anode terminal and the cathodeterminal. To provide optical-to-electrical conversion of the digital bitstream, the heterojunction thyristor device switches from its OFF stateto its ON state in the event that, during a given sampling period, thelight intensity level of the input digital optical signal corresponds tothe ON logic level; however, it does not switch into the ON state (andremains in the OFF state) in the event that, during the given samplingperiod, the light intensity level of the digital optical signalcorresponds to the OFF logic level.

When using electrical sampling pulses as the sampling clock, thesesampling pulses can be in the form of downward running electrical pulses(e.g., pulses wherein the relative voltage between the start of thepulse and the peak of the pulse is less than zero) supplied to then-channel injector terminal(s) of the heterojunction thyristor device,and/or in the form of upward running electrical pulses (e.g., pulseswherein the relative voltage between the start of the pulse and the peakof the pulse is greater than zero) supplied to the p-channel injectorterminal(s) of the heterojunction thyristor device.

The voltage level (e.g., magnitude) of the ON state of the outputdigital electrical signal produced by the heterojunction thyristordevice can be adjusted by a voltage divider network coupled between thecathode terminal of the device and ground potential.

A plurality of such heterojunction thyristor devices may be configuredto convert a plurality of synchronous digital optical signals tocorresponding digital electrical signals for use in high speedapplications, such as a receive module in a parallel optical data link.

In another aspect of the present invention, a plurality ofheterojunction thyristor devices are configured to convert a digitalword encoded by a parallel digital optical signal (e.g., a plurality ofsynchronous optical bits) to an output analog electrical signal whosemagnitude corresponds to the digital word. Each heterojunction thyristordevice is configured to convert an optical bit in the digital word to acorresponding digital electrical signal. The voltage levels (e.g.,magnitudes) of the ON state of the digital electrical signals producedby the heterojunction thyristor devices are varied by voltage dividernetworks coupled between the cathode terminal of the devices and groundpotential. In this manner, the voltage divider networks produceelectrical signals whose magnitude corresponds to the contribution ofeach optical bit in the digital word. The electrical signals produced bythe voltage divider networks is summed by a summing network to generatethe output analog electrical signal corresponding to the digital word.

Preferably, the summing network includes a chain of two-port addingnodes and sample/hold circuits arranged as pairs, each corresponding toa different voltage divider network. In this configuration, the outputelectrical signal generated by a given voltage divider network issupplied to an input node of the two-port adding node of thecorresponding pair.

In another aspect of the present invention, a heterojunction thyristordevice is configured as an optically-controlled (orelectrically-controlled) sampling/switching device. In thisconfiguration, first and second channel regions are disposed between theanode terminal and the cathode terminal of the device, and an electricalinput terminal and an electrical output terminal are coupled to oppositeends of the first channel region. At least one optical control signal(or an electrical control signal) is supplied to the device. When thelight intensity level of the at least one optical control signal (ormagnitude of the electrical control signal) corresponds to apredetermined ON condition, sufficient charge is stored in the secondchannel region to cause the heterojunction thyristor device to operatein an ON state whereby current flows between the anode terminal and thecathode terminal and the electrical input terminal is electricallycoupled to the electrical output terminal. When the light intensitylevel of the at least one optical control signal (or magnitude of theelectrical control signal) corresponds to a predetermined OFF condition,the heterojunction thyristor device operates in an OFF state wherebycurrent does not flow between the anode terminal and the cathodeterminal and the electrical input terminal is electrically isolated fromthe electrical output terminal.

The optical control signal can be an optical sampling clock, a digitaloptical signal encoding bits of information, or the combination of adigital optical signal and an optical sampling clock (which definessampling periods that overlap the bits of information in the digitaloptical signal). The electrical control signal can be an electricalsampling clock injected into the second channel of the device thatoperates alone to supply charge that induces ON state operation.Alternatively, in addition to the optical control signal(s), anelectrical sampling clock can be injected into the second channel of thedevice to contribute to the supply of charge therein that induces ONstate operation (when the light intensity level of the optical controlsignal corresponds to the predetermined ON condition).

Importantly, such optically-controlled (or electrically-controlled)sampling/switching devices are suitable for use as sample and holdcircuitry in the photonic digital-to-analog converters described herein,and in a wide variety of signal processing applications, such asanalog-to-digital converters, switched-capacitor filters, andswitched-capacitor waveform shaping circuits.

In another aspect of the present invention, a plurality ofheterojunction thyristor devices are configured to convert a digitalword encoded by a parallel digital optical signal (e.g., a plurality ofsynchronous optical bits) to an output analog electrical signal whosemagnitude corresponds to the digital word. Each heterojunction thyristordevice is configured as a sampling device to convert an optical bit inthe digital word to a corresponding digital electrical signal. Thevoltage levels (e.g., magnitudes) of the ON state of the digitalelectrical signals produced by the heterojunction thyristor devices areprovided by reference voltage sources operably coupled to the inputterminals of the heterojunction thyristor devices. In this manner, theheterojunction thyristor devices and corresponding voltage referencesources produce electrical signals whose magnitude corresponds to thecontribution of each optical bit in the digital word. These electricalsignals are summed by a summing circuit, which is preferably implementedby another heterojunction-thyristor-based sampling device, to generatethe output analog electrical signal corresponding to the digital word.

In another aspect of the present invention, a heterojunction thyristordevice configured for optically-controlled sampling/switching is used asthe basis for converting a digital word encoded by a serial digitaloptical data signal (e.g., serial optical bit stream) into acorresponding analog electrical signal. In this configuration, a voltagereference is operably coupled to the electrical input terminal of theheterojunction thyristor device. The voltage reference cooperates withthe heterojunction thyristor device to sequentially generate at itselectrical output terminal a voltage signal representing thecontribution of each bit of the digital word encoded in the serialdigital optical data signal. A summing network is operably coupled tothe electrical output terminal of the device. The summing networksequentially sums the voltage signal over the sequence of bits toproduce an analog electrical signal corresponding to the digital wordfor output therefrom.

Preferably, the summing network includes an adding node, sample and holdcircuit, and a feedback path between the sample and hold circuit and theadding node. In addition, the voltage reference preferably supplies avoltage level corresponding to the maximum voltage level of the analogelectrical signal divided by 2^((N−1)), where N is the number of bits insaid digital word, and the feedback path comprises an amplifier thatamplifies the output of the sample and hold circuit by a factor of 2.

According to other embodiments of the present invention, monolithicoptoelectronic integrated circuits that include one or moreheterojunction thyristor devices as described herein are integrated withelectronic devices (such as transistors) and/or optical devices (such aswaveguide devices).

Additional objects and advantages of the invention will become apparentto those skilled in the art upon reference to the detailed descriptiontaken in conjunction with the provided figures.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A is a functional block diagram illustrating a prior art mechanismthat converts a digital optical signal to a digital electrical signal.

FIG. 1B is a functional block diagram illustrating a prior art paralleloptical data link.

FIG. 1C is a functional block diagram illustrating a prior art mechanismthat converts a digital optical signal to an analog electrical signal.

FIG. 2A is a cross-sectional schematic showing a layer structure inaccordance with the present invention, and from which devices of thepresent invention can be made.

FIG. 2B1 is a pictorial illustration of a heterojunction thyristordevice that is used to convert an input digital optical signal to anoutput digital electrical signal in accordance with the presentinvention; in this configuration, the input digital optical signal ispart of the Optical IN signal that is resonantly absorbed by the device,and a sampling clock (electrical) is supplied to the injector terminalof the device.

FIG. 2B2 is a pictorial illustration of a heterojunction thyristordevice that is used to convert an input digital optical signal to anoutput digital electrical signal in accordance with the presentinvention; in this configuration, the input digital optical signal and asampling clock (optical) are part of the Optical IN signal that isresonantly absorbed by the device.

FIG. 2B3 is a pictorial illustration of a heterojunction thyristordevice that is used to convert an input digital optical signal to anoutput digital electrical signal in accordance with the presentinvention; a voltage divider network is used to adjust magnitude of theON state of the digital electrical signal; in this configuration, theinput digital optical signal is part of the Optical IN signal that isresonantly absorbed by the device, and a sampling clock (electrical) issupplied to the injector terminal of the device.

FIG. 2B4 is a pictorial illustration of a heterojunction thyristordevice that is used to convert an input digital optical signal to anoutput digital electrical signal in accordance with the presentinvention; a voltage divider network is used to adjust magnitude of theON state of the digital electrical signal; in this configuration, theinput digital optical signal and a sampling clock (optical) are part ofthe Optical IN signal that is resonantly absorbed by the device.

FIG. 2C is a graph showing the current-voltage characteristics of theheterojunction thyristor device of the present invention over varyinginjector currents (I_(g)), and the bias line that depicts operation ofthe heterojunction thyristor device in converting the input digitaloptical signal to an output digital electrical signal.

FIG. 2D1 is a graph depicting the operation of the heterojunctionthyristor device in converting an input digital optical signal (in theON state) to a corresponding output digital electrical signal (in the ONstate) during a sampling period defined by an electrical sampling clocksupplied to the injector terminal of the device; conversely, when theinput digital optical signal is in the OFF state, the heterojunctionthyristor remains in its NON—CONDUCTING/OFF state and produces an outputdigital electrical signal in the OFF state during the correspondingsampling period defined by the electrical sampling clock.

FIG. 2D2 is a graph depicting the operation of the heterojunctionthyristor device in converting an input digital optical signal (in theON state) to a corresponding output digital electrical signal (in the ONstate) during a sampling period defined by an optical sampling clock;conversely, when the input digital optical signal is in the OFF state,the heterojunction thyristor remains in its NON-CONDUCTING/OFF state andproduces an output digital electrical signal in the OFF state during thecorresponding sampling period defined by the optical sampling clock.

FIG. 3A is a schematic showing an exemplary layer structure made withgroup III-V material in accordance with the present invention, and fromwhich devices of the present invention can be made.

FIG. 3B shows the energy band diagram of the structure of FIG. 3A.

FIG. 3C is a cross-sectional schematic view showing the generalizedconstruction of an exemplary heterojunction thyristor formed from thelayer structure of FIG. 3A.

FIG. 3D is a schematic showing an alternate layer structure made withgroup III-V material in accordance with the present invention, and fromwhich devices of the present invention can be made.

FIG. 3E shows the energy band diagram of the structure of FIG. 3D.

FIG. 3F is a cross-sectional schematic view showing the generalizedconstruction of an exemplary heterojunction thyristor formed from thelayer structure of FIG. 3D.

FIG. 4A is a functional block diagram illustrating a photonicdigital-to-analog converter that converts a digital word encoded by aparallel digital optical signal (e.g., a plurality of synchronousoptical bits) to an output analog electrical signal whose magnitudecorresponds to the digital word in accordance with the presentinvention; in this configuration, electrical clock signals are used toperform the conversion operations.

FIGS. 4B(i), 4B(ii), and 4B(iii) is a timing diagram illustrating theencoding of a 4-bit digital word by 4 input digital optical signals andassociated electrical timing signals relative thereto; FIG. 4B(i) is asignal diagram that illustrates the encoding of a 4-bit digital word by4 input digital optical signals supplied to the heterojunction thyristordevices of FIG. 4A; FIG. 4B(ii) is a signal diagram that illustrates anelectrical timing signal A supplied to the injector terminals of theheterojunction thyristor devices of FIG. 4A; and FIG. 4B(iii) is asignal diagram that illustrates an electrical timing signal B suppliedto the summing network of FIG. 4A.

FIGS. 4C and 4D are cross-sectional schematics showing heterojunctionthyristor devices configured to perform electrically-controlled samplingoperations in response to an electrical clock signal supplied thereto;each of these configurations is suitable to realize theelectrically-controlled sample and hold circuit of FIG. 4A.

FIG. 4E is a functional block diagram illustrating a photonicdigital-to-analog converter that converts a digital word encoded by aparallel digital optical signal (e.g., a plurality of synchronousoptical bits) to an output analog electrical signal whose magnitudecorresponds to the digital word in accordance with the presentinvention; in this configuration, electrical clock signals are used toperform the conversion operations.

FIG. 5A is a functional block diagram illustrating a photonicdigital-to-analog converter that converts a digital word encoded by aparallel digital optical signal (e.g., a plurality of synchronousoptical bits) to an output analog electrical signal whose magnitudecorresponds to the digital word in accordance with the presentinvention; in this configuration, optical clock signals are used toperform the conversion operations.

FIGS. 5B(i), 5B(ii), and 5B(iii) is a timing diagram illustrating theencoding of a 4-bit digital word by 4 input digital optical signals andassociated optical timing signals relative thereto; FIG. 5B(i) is asignal diagram that illustrates the encoding of a 4-bit digital word bythe input digital optical signals supplied to the heterojunctionthyristor devices of FIG. 5A; FIG. 5B(ii) is a signal diagram thatillustrates an optical timing signal A supplied to the heterojunctionthyristor devices of FIG. 5A for resonant absorption therein; and FIG.5B(iii) is a signal diagram that illustrates an optical timing signal Bsupplied to the summing network of FIG. 5A.

FIGS. 5C and 5D are cross-sectional schematics showing heterojunctionthyristor devices configured to perform optically-controlled samplingoperations in response to an optical clock signal supplied thereto; eachof these configurations is suitable to realize the optically-controlledsample and hold circuit of FIG. 5A.

FIG. 5E is a functional block diagram illustrating a photonicdigital-to-analog converter that converts a digital word encoded by aparallel digital optical signal (e.g., a plurality of synchronousoptical bits) to an output analog electrical signal whose magnitudecorresponds to the digital word in accordance with the presentinvention; in this configuration, optical clock signals are used toperform the conversion operations.

FIG. 5F is a cross-sectional schematic view showing the generalizedconstruction of an exemplary heterojunction thyristor device formed fromthe layer structure of FIG. 3A, which is readily configured inaccordance with FIGS. 4C-4E or 5C-5E to perform sampling operations.

FIG. 6A is a functional block diagram illustrating a photonicdigital-to-analog converter that converts a digital word encoded by aserial digital optical signal (e.g., a plurality of serial optical bits)to an output analog electrical signal whose magnitude corresponds to thedigital word in accordance with the present invention; in thisconfiguration, an electrical clock signal is used to perform theconversion operations.

FIGS. 6B(i) and 6B(ii) is a timing diagram illustrating the serialencoding of a 4-bit digital word by the input digital optical signal andassociated electrical clock signal relative thereto; FIG. 6B(i) is asignal diagram that illustrates the serial encoding of a 4-bit digitalword by the input digital optical signal supplied to the heterojunctionthyristor device of FIG. 6A; and FIG. 6B(ii) is a signal diagram thatillustrates the electrical clock signal supplied to the injectorterminal of the first heterojunction thyristor device and supplied tothe summing network of FIG. 6A.

FIGS. 6C and 6D are cross-sectional schematics showing heterojunctionthyristor devices configured to perform optically-controlled samplingoperations in response to a digital optical signal and an electricalclock signal supplied thereto; each of these configurations is suitableto realize the first heterojunction sampling device of FIG. 6A.

FIG. 7A is a functional block diagram illustrating a photonicdigital-to-analog converter that converts a digital word encoded by aserial digital optical signal (e.g., a plurality of serial optical bits)to an output analog electrical signal whose magnitude corresponds to thedigital word in accordance with the present invention; in thisconfiguration, an optical clock signal is used to perform the conversionoperations.

FIGS. 7B(i) and 7B(ii) is a timing diagram illustrating the serialencoding of a 4-bit digital word by the input digital optical signal andassociated optical clock signal relative thereto; FIG. 7B(i) is a signaldiagram that illustrates the serial encoding of a 4-bit digital word bythe input digital optical signal supplied to the heterojunctionthyristor device of FIG. 7A; and FIG. 7B(ii) is a signal diagram thatillustrates the optical clock signal supplied to the firstheterojunction thyristor device and supplied to the summing network ofFIG. 7A.

FIGS. 7C and 7D are cross-sectional schematics showing heterojunctionthyristor devices configured to perform optically-controlled samplingoperations in response to a digital optical signal and an optical clocksignal supplied thereto; each of these configurations is suitable torealize the first heterojunction sampling device of FIG. 7A.

FIG. 8A is a cross-sectional schematic view showing the generalizedconstruction of an exemplary p-type quantum-well-base bipolar transistordevice formed from the layer structure of FIG. 2A.

FIG. 8B is a diagram illustrating a differential amplifier circuithaving a gain factor of 2, which is realized with a plurality of p-typequantum-well-base bipolar transistor devices as shown in FIG. 8A; thisconfiguration is suitable for the feedback amplifier in the summingnetwork of FIGS. 6A and 7A.

FIG. 9A is a functional block diagram illustrating anoptical-to-electrical converter that converts a digital optical signal(e.g., serial optical bit stream) to a digital electrical signal; inthis configuration, electrical clock signals are used to perform theconversion operations.

FIGS. 9B(i) and 9B(ii) is a timing diagram illustrating the serialencoding of a 4-bit digital word by the input digital optical signal andassociated electrical timing signals relative thereto; FIG. 9B(i) is asignal diagram that illustrates the serial encoding of bits in thedigital optical signal supplied to the heterojunction thyristor deviceof FIG. 9A; and FIG. 9B(ii) is a signal diagram that illustrates theelectrical timing signal A supplied to the injector terminal of theheterojunction thyristor device of FIG. 9A.

FIG. 9C is a functional block diagram illustrating anoptical-to-electrical converter that converts a digital optical signal(e.g., serial optical bit stream) to a digital electrical signal; inthis configuration, optical clock signals are used to perform theconversion operations.

FIGS. 9D(i) and 9D(ii) is a timing diagram illustrating the serialencoding of a 4-bit digital word by the input digital optical signal andassociated optical timing signals relative thereto; FIG. 9D(i) is asignal diagram that illustrates the serial encoding of bits in thedigital optical signal supplied to the heterojunction thyristor deviceof FIG. 9C; and FIG. 9D(ii) is a signal diagram that illustrates theoptical clock signal A supplied to the heterojunction thyristor deviceof FIG. 9C.

FIG. 9E is a functional block diagram of a receive module of a paralleldata link, the receive module employing a plurality of theoptical-to-electrical converters of FIG. 9A (or FIG. 9C) to convert aplurality of synchronous optical bit streams to corresponding digitalelectrical signals.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Modulation-doped quantum well heterojunction transistors—including wellknown Pseudomorphic Pulsed Doped High Electron Mobility Transistors(Pulsed Doped PHEMT), which are sometimes referred to as Pulsed DopedModulation Doped Field Effect Transistors (Pulsed Doped MODFET) orPulsed Doped Two Dimensional Gas Field Effect Transistors (Pulsed DopedTEGFET)— have become well recognized for their superior low noise andhigh frequency performance and are now in demand in many high frequencyapplications (e.g., front end amplifier in wireless communicationssystems and in Monolithic Microwave and Millimeterwave IC (MMIC)designs). GaAs/InGaAs/Al_(x)Ga_(1-x)As or InP/InGaAs/InAlAs is the III-Vmaterial system of choice for these devices because of the ability togrow high optical/electrical quality epitaxial layers by molecular beamepitaxy (MBE). Alternatively, strained silicon heterostructuresemploying silicon-germanium (SiGe) layers have been used to produce suchdevices.

U.S. Pat. No. 4,827,320 to Morkoc et al. discloses a pseudomorphic HEMT(PHEMT) structure that employs a layer of strained InGaAs (undoped)between a GaAs substrate and a layer of undoped AlGaAs to form a quantumwell defined by the strained InGaAs layer. A layer of n+ doped AlGaAs isformed on the undoped AlGaAs layer. A layer of n+ GaAs is formed on thelayer of n+ doped AlGaAs. The layer of n+ GaAs facilitates an ohmiccontact to source/drain electrodes. A gate electrode of aluminum isrecessed below the layer of n+ GaAs and a portion of the n+ AlGaAs layerby wet chemical etch and evaporation of aluminum.

The PHEMT structure has been very successful in producing microwavetransistors that operate well into the multi-gigahertz regime, initiallybeing used extensively in military systems and now finding their wayinto commercial products, particularly in the area of cellularcommunications. In recent years, there has been a growing interest incombining the PHEMT with optical capability because of the difficulty inpropagating very high frequency signals to and from the integratedcircuit by coaxial lines. Combining electronic with optoelectroniccomponents monolithically gives rise to the concept of theoptoelectronic integrated circuit (OEIC). However, there are seriousproblems encountered because of the dissimilar nature of the structuresof the FET, the pn junction laser, PIN diode, etc.

To achieve this goal, inversion channel heterojunction structurescreated from a single epitaxial growth have been used to realize a rangeof optoelectronic devices including lasers, detectors and field effecttransistors (FETs). An exemplary inversion channel heterojunctionstructure is described in Taylor and Kiely, “Theoretical andExperimental Results for the Inversion Channel Heterostructure FieldEffect Transistors”, IEE Proceedings-G, Vol. 140, No. 6, Dec. 1993. Inthis structure, for the region between the modulation doping layer andthe gate of the semiconductor surface, the doping of this region issubstantially p type in order to provide a low resistance ohmic contactfor the gate of the FET. However, the high p-type doping of this regioncreates many problems, including:

-   -   i) the effects of free carrier absorption makes formation of a        vertical cavity laser difficult;    -   ii) forming depletion-type FETs by implanting n-type dopant is        difficult; this difficulty stems from the difficulty in        controlling the dopant density in the bulk region; more        specifically, compensating a large p density with a large n        density to obtain a lower p density is difficult to control in a        bulk region (but much easier in a delta doped region);    -   iii) controlling the threshold voltage of an enhancement type        FET is difficult because the input capacitance is a function of        doping which is harder to control than layer thickness; and    -   iv) producing effective current funneling for inducing lasing is        difficult; more specifically, it is very desirable to create a        pn junction by N type implantation to steer the current in this        structure since this would be compatible with the overall        approach to building the FET devices; the heavy p doping bulk        layers makes it difficult to create junction isolation that has        low leakage.

The present invention builds upon novel device structures utilizingmodulation-doped quantum well heterojunctions that do not suffer fromthe problems associated with the prior art PHEMT devices. Such noveldevice structures are described in detail in U.S. Pat. No. 6,031,243;U.S. patent application Ser. No. 09/556,285 (Attorney Docket No.OPE-002), filed on Apr. 24, 2000; U.S. patent application Ser. No.09/798,316 (Attorney Docket No. OPE-004), filed on Mar. 2, 2001; U.S.patent application Ser. No. 08/949,504 (Attorney Docket No. OPE-005),filed on Oct. 14, 1997, U.S. patent application Ser. No. 10/200,967(Attorney Docket No. OPE-005-CIP), filed on Jul. 23, 2002; U.S.application Ser. No. 09/710,217 (Attorney Docket No. OPE-006), filed onNov. 10, 2000; U.S. patent application Ser. No. 60/376,238 (AttorneyDocket No. OPE-008-PROV), filed on Apr. 26, 2002; U.S. patentapplication Ser. No. 10/280,892 (Attorney Docket No. OPE-012), filed onOct. 25, 2002; each of these references herein incorporated by referencein its entirety.

The general structure of the heterojunction thyristor device isillustrated in the cross-section of FIG. 2A. In addition, the generalstructure of FIG. 2A can be configured to operate as a field effecttransistor, bipolar transistor, and as a passive waveguide such thatthese devices can be integrated to form a monolithic optoelectronicintegrated circuit as described herein.

Turning now to FIG. 2A, the heterojunction thyristor device 1 of thepresent invention includes a bottom dielectric distributed braggreflector (DBR) mirror 12 formed on substrate 10. The bottom DBR mirror12 typically is formed by depositing pairs of semiconductor ordielectric materials with different refractive indices. When twomaterials with different refractive indices are placed together to forma junction, light will be reflected at the junction. The amount of lightreflected at one such boundary is not necessarily large. However, ifmultiple junctions/layer pairs are stacked periodically with each layerhaving a quarter-wave (¼n) optical thickness, the reflections from eachof the boundaries will be added in phase to produce a large amount ofreflected light (e.g., a large reflection coefficient) at the particularcenter wavelength I_(D). Deposited upon the bottom DBR mirror 12 is theactive device structure which consists of two HFET devices. The first ofthese is a p-channel HFET 11 (comprising layers 14,16,18,20 and 22)which has a p-type modulation doped quantum well and is positioned withthe gate terminal on the lower side (i.e. on the mirror as justdescribed) and the collector terminal on the upper side. The second ofthese is an n-channel HFET 13 (comprising layers 22,24,26,28,30) whichhas an n-type modulation doped quantum well and is positioned with thegate terminal on the top side and the collector terminal on the lowerside which is the collector of the p-channel device. Therefore anon-inverted N-channel device is stacked upon an inverted p-channeldevice to form the active device structure.

The active device layer structure begins with n-type ohmic contactlayer(s) 14 which enables the formation of ohmic contacts thereto. Asshown, ohmic contact layer 14 is operably coupled to cathode terminal 40of the heterojunction thyristor device (which corresponds to the gateelectrode of the p-channel HFET device). Deposited on layer 14 is one ormore n-type layers 16 and an undoped spacer layer 18 which serveelectrically as part of the P-channel HFET gate and optically as lowerwaveguide cladding layers. Deposited on layer 18 is a p-type modulationdoped heterojunction structure 20 that defines one or more quantum wells(which may be formed from strained or unstrained heterojunctionmaterials). Deposited on p-type modulation doped heterojunctionstructure 20 is an undoped spacer layer 22, which forms the collector ofthe P-channel HFET device. All of the layers grown thus far form theP-channel HFET device with the gate ohmic contact on the bottom.

Undoped spacer layer 22 also forms the collector region of the N-channelHFET device. Deposited on layer 22 is a n-type modulation dopedheterojunction structure 24 that defines one or more quantum wells(which may be formed from strained or unstrained heterojunctionmaterials). Deposited on the n-type modulation doped heterojunctionstructure 24 is an undoped spacer layer 26 and one or more p-type layers28 which serve electrically as part of the n-channel HFET gate andoptically as upper waveguide cladding layers. Preferably, the p-typelayers 28 include two sheets of planar doping of highly doped p-materialseparated by a lightly doped layer of p-material. These p-type layersare separated from the N-type modulation doped quantum well (QW)heterostructure 24 by undoped spacer material 26. In this configuration,the top charge sheet achieves low gate contact resistance and the bottomcharge sheet defines the capacitance of the n-channel HFET with respectto the N-type modulation doped QW heterostructure 24. Deposited onp-type layer(s) 28 is a p-type ohmic contact layer(s) 30 which enablesthe formation of ohmic contacts thereto. As shown, ohmic contactlayer(s) 30 is operably coupled to the anode terminal 36 of theheterojunction thyristor device (which corresponds to the gate electrodeof the n-channel HFET device).

The injector terminal 38 of the heterojunction thyristor device (whichis analogous to the gate terminal of conventional thyristor devices)preferably is operably coupled to the QW channel(s) realized in theN-type modulation doped QW(s) heterostructure 24 as shown in FIG. 2A.Alternatively, the injector terminal of the heterojunction thyristordevice may be operably coupled to the QW channel(s) realized in theP-type modulation doped QW(s) heterostructure 20. In such aconfiguration, the polarity of the control signals applied to theinjector terminal 38 are reversed, and the bias current is configured tomove charge from the QW channel(s) realized in the P-type modulationdoped QW(s) heterostructure 24 to ground potential.

As another alternative, a first injector terminal may be operablycoupled to the QW channel(s) realized in the N-type modulation dopedQW(s) heterostructure 24 while a second injector terminal is operablycoupled to the P-type modulation doped QW(s) heterostructure 20. In sucha configuration, the polarity of the control signals applied to thesecond injector terminal are reversed, and the bias current isconfigured to move charge from the QW channel(s) realized in the P-typemodulation doped QW(s) heterostructure 20 to ground potential.

To form a resonant cavity device where light is input into the devicelaterally (i.e., from a direction normal to the cross section of FIG.2A), a diffraction grating and top DBR mirror are formed over the activedevice structure described above. In this configuration, the diffractiongrating performs the function of diffracting incident light that ispropagating in the lateral direction into the vertical cavity mode,where it is absorbed resonantly in the vertical cavity. Alternatively,light may enter the resonant vertical cavity through an optical aperture(not shown) in the top surface of the device. In this case, thediffraction grating is omitted, the top DBR mirror defines a cavity forthe vertical absorption of light, and the device operates as a verticalcavity detector. In either configuration, the distance between the topDBR mirror and bottom DBR mirror preferably represents an integralnumber of ¼ wavelengths at the designated wavelength. This distance iscontrolled by adjusting the thickness of one or more of the layerstherebetween to enable this condition. These configurations define anoptically active region 44 that encompasses the QW channel(s) ofstructures 24 and 20. The optical signal IN 42 (which propagates in thevertical direction, or which propagates in the lateral direction and isdiffracted from the lateral direction into a vertical propagationdirection by diffraction grating 32) is resonantly absorbed in region44. Such absorption results in the generation of electron-hole pairs inthe QW channel(s) thereby causing charge to build up therein, which maybe sufficient to induce a change in current flowing through the devicefrom the anode terminal 36 to the cathode terminal 40.

The operation of the heterojunction thyristor device can also becontrolled by injection of electrical energy (e.g., an electrical inputpulse signal) into the QW channel(s) of structure 24 (and/or the QWchannel(s) of structure 20). Such injection contributes to thegeneration of electron-hole pairs in the QW channel(s) thereby causingcharge to build up therein.

FIGS. 2B1 through 2B4, 2C, 2D1 and 2D2 illustrate the operationalcharacteristics of the heterojunction thyristor device of FIG. 2A inaccordance with the present invention. The device switches from anon-conducting/OFF state (where the current I is substantially zero) toa conducting/ON state (where current I is substantially greater thanzero) when: i) the anode terminal 36 is forward biased (e.g. biasedpositively) with respect to the cathode terminal 40; and ii) absorptionof optical energy in the QW channel(s) of N-type modulation dopedheterojunction QW structure 24 and/or injection of electrical energyinto the QW channel(s) of N-type modulation doped heterojunction QWstructure 24 produce a charge in the N-type modulation doped QWstructure 24 that is greater than the critical switching charge Q_(CR),which is that charge that reduces the forward breakdown voltage suchthat no off state bias point exists. The critical switching chargeQ_(CR) is unique to the geometries and doping levels of the device. Theforward breakdown voltage of the device varies over the injector currentI_(g) as shown in FIG. 2C.

The device switches from the conducting/ON state to a non-conducting/OFFstate when the current I through the device falls below the hold currentI_(H) of the device for a sufficient period of time such that the chargein the N-type modulation doped QW structure 24 decreases below theholding charge Q_(H). The holding charge Q_(H) is the critical value ofthe channel charge which will sustain holding action.

The device is configured to convert an input digital optical signal toan output digital electrical signal as follows. The input digitaloptical signal encodes bits of information (each bit representing an OFFlogic level or ON logic level) and is part of the Optical IN signal thatis resonantly absorbed by the device. A sampling clock defines samplingperiods that overlap the bits (e.g., ON/OFF pulse durations) in theinput digital optical signal. In FIGS. 2B1 and 2B3, the sampling clockis in the form of downward running electrical pulses (e.g., pulseswherein the relative voltage between the start of the pulse and the peakof the pulse is less than zero) supplied to the injector terminal 38.Alternatively, as shown in FIGS. 2B2 and 2B4, the sampling clock is inthe form of optical pulses that are part of the Optical IN signal thatis resonantly absorbed by the device. In addition, the device is biased(preferably, by selecting the appropriate load resistance/voltagedivider network as shown in FIGS. 2B1 through 2B4) such that the currentI through the device in the ON state is substantially greater than zerobut substantially below the threshold current for lasing I_(L)(preferably about one-third of IL) as shown in FIG. 2C. In addition, theinjector terminal 38 is forward biased with respect to the anodeterminal 36 through a current source that generates a bias currentI_(BIAS) as shown.

In the configurations of FIGS. 2B1 and 2B3, the length and width of thedevice is sized such that it operates during a given sampling perioddefined by the electrical sampling clock as follows. When the lightlevel of the input digital optical signal corresponds to the ON logiclevel, channel current produced by the combination of the input digitaloptical signal and the electrical sampling pulse exceeds the biascurrent I_(BIAS) to produce the critical switching charge Q_(CR) in theN-type modulation doped QW structure 24. This causes the heterojunctionthyristor to switch to its conducting/ON state where the current Ithrough the device is substantially greater than zero but below thethreshold for lasing I_(L). This operation is shown graphically in FIG.2D1. However, when the light level of the input digital optical signalfalls to the OFF logic level, the bias current I_(BIAS) exceeds thechannel current produced by the electrical sampling clock alone and thusdraws on the injector terminal 38 to drain charge from the N-typemodulation doped QW structure 24, which causes the channel charge tofall below the holding charge Q_(H). This causes the heterojunctionthyristor to switch to its non-conducting/OFF state where the current Ithrough the device is substantially zero. When the light level of theinput digital optical signal corresponds to the OFF logic level, thebias current I_(BIAS) exceeds the combination of the channel currentproduced by the input digital optical signal and the electrical samplingclock and thus draws on the injector terminal 38 to drain charge fromthe N-type modulation doped QW structure 24, which causes the channelcharge to remain below the holding charge Q_(H). This causes the deviceto remain in its non-conducting/OFF state where the current I throughthe device is substantially zero. In this manner, the logic levels (ONstate/OFF state) of the output digital electrical signal produced at thecathode terminal 50 correspond to the logic levels of the input digitaloptical signal. The length and width of the device must also be selectedsuch that the output current produced at the cathode terminal 40 islarge enough to drive the capacitive load of the circuit element(s)coupled thereto, and leakage currents at the periphery of the devicedoes not degrade the signal to noise ratio of the device to unacceptablelevels.

The voltage level (e.g., magnitude) of the ON state of the outputdigital electrical signal produced by the device can be adjusted by avoltage divider network coupled between the cathode terminal 40 andground potential as shown in FIG. 2B3.

In the configurations of FIGS. 2B2 and 2B4, the length and width of thedevice are sized such that it operates during a given sampling perioddefined by the optical sampling clock as follows. When the light levelof the input digital optical signal corresponds to the ON logic level,channel current produced by the combination of the input digital opticalsignal and the optical sampling pulse exceeds the bias current I_(BIAS)to produce the critical switching charge Q_(CR) in the N-type modulationdoped QW structure 24. This causes the heterojunction thyristor toswitch to its conducting/ON state where the current I through the deviceis substantially greater than zero but below the threshold for lasingI_(L). This operation is shown graphically in FIG. 2D2. However, whenthe light level of the input digital optical signal falls to the OFFlogic level, the bias current I_(BIAS) exceeds the channel currentproduced by the optical sampling pulse alone and thus draws on theinjector terminal 38 to drain charge from the N-type modulation doped QWstructure 24, which causes the channel charge to fall below the holdingcharge Q_(H). This causes the heterojunction thyristor to switch to itsnon-conducting/OFF state where the current I through the device issubstantially zero. When the light level of the input digital opticalsignal corresponds to the OFF logic level, the bias current I_(BIAS)exceeds the combination of the channel current produced by the inputdigital optical signal and the optical sampling clock and thus draws onthe injector terminal 38 to drain charge from the N-type modulationdoped QW structure 24, which causes the channel charge to remain belowthe holding charge Q_(H). This causes the device to remain in itsnon-conducting/OFF state where the current I through the device issubstantially zero. In this manner, the logic levels (ON state/OFFstate) of the output digital electrical signal produced at the cathodeterminal 40 correspond to the logic levels of the input digital opticalsignal. The length and width of the device must also be selected suchthat the output current produced at the cathode terminal 40 is largeenough to drive the capacitive load of the circuit element(s) coupledthereto, and leakage currents at the periphery of the device does notdegrade the signal to noise ratio of the device to unacceptable levels.

The voltage level (e.g., magnitude) of the ON state of the outputdigital electrical signal produced by the device can be adjusted by avoltage divider network coupled between the cathode terminal 40 andground potential as shown in FIG. 2B4.

As previously mentioned, in an alternate embodiment, the injectorterminal 38 of the device may be operably coupled to the p-type QWchannel(s) realized in the P-type modulation doped QW(s) structure 20.In such a configuration, the electrical sampling clock of FIGS. 2B1 and2B3 is in the form of upward running electrical pulses (e.g., pulseswherein the relative voltage between the start of the pulse and the peakof the pulse is greater than zero) supplied to the injector terminal ofthe device. In addition, the bias current source of FIGS. 2B1 through2B4 draws charge from the p-type QW channel(s) to ground potential.

Also, as previously mentioned, in another alternative embodiment, afirst injector terminal may be operably coupled to the n-type QWchannel(s) realized in the N-type modulation doped QW structure 24 whilea second injector terminal is operably coupled to the p-type QWchannel(s) in the P-type modulation doped QW(s) structure 20. In such aconfiguration, the electrical sampling clock supplied to the p-type QWchannel(s) is in the form of upward running electrical pulses (e.g.,pulses wherein the relative voltage between the start of the pulse andthe peak of the pulse is greater than zero) supplied to the secondinjector terminal of the device. In addition, the bias current sourceoperably coupled to the p-type QW channel(s) draws charge from thep-type QW channel(s) to ground potential.

The structure of FIG. 2A may also be used to produce various transistordevices, including n-channel HFET devices, p-channel HFET devices,p-type quantum-well-base bipolar transistor devices and n-typequantum-well-base bipolar transistor devices.

In a n-channel HFET, ohmic metal source and drain electrodes areelectrically coupled to spaced apart N-type implants, which areelectrically coupled to the n-type QW structure 24 to form a channelregion there between. An ohmic metal gate electrode is formed on thep-type ohmic contact layer 30 and covers the channel region. An ohmicmetal collector electrode is electrically coupled to at least one P-typeimplant, which is electrically coupled to the p-type QW structure 20below the channel region.

In a p-channel HFET, ohmic metal source and drain electrodes areelectrically coupled to spaced apart p-type implants, which areelectrically coupled to the p-type QW structure 20 to form a channelregion there between. Outside the channel region, an ohmic metal gateelectrode is deposited on the n-type ohmic contact layer 14. An n-typeimplant is deposited above layer 22 and preferably into layer 24. Anohmic metal collector electrode is formed on the n-type implant.

In a p-type quantum-well-base bipolar transistor device, one or morebase electrodes are electrically coupled to spaced apart P-typeimplants, which are electrically coupled to the p-type QW structure 20.Outside the p-type implants, one or more emitter electrodes aredeposited on the n-type ohmic contact layer 14. A collector electrode iselectrically coupled to an n-type implant, which is electrically coupledto the n-type QW structure 24. An additional collector electrode may beelectrically coupled to another n-type implant into the p-type materialof layer 28 or into the undoped spacer 26.

In an n-type quantum-well-base bipolar transistor device, one or morebase electrodes are electrically coupled to spaced apart n-typeimplants, which are electrically coupled to the n-type QW structure 24.One or more collector electrodes are electrically coupled tocorresponding p-type implants, which are electrically coupled to thep-type QW structure 20. An emitter electrode is deposited on the n-typeohmic contact layer 30.

In addition, the structure of FIG. 2A may be used to produce variousoptoelectronic components, such as a laser device or an in-plane passivewaveguide. To configure the structure as a laser, the heterojunctionthyristor device is biased (preferably, by selection of load resistanceoperably coupled between the cathode terminal 40 and ground potential)such that the current I flowing the through the device in theconducting/ON state is above the lasing threshold I_(L) shown in FIG.2C. The conducting-ON state is controlled by absorption of an opticalcontrol signal incident on the device (and/or by injection of anelectrical control signal supplied to the injector terminal 38) whichcauses charge to build up in the QW channel(s) of the device sufficientto induce a change in current flowing through the device from the anodeterminal 36 to the cathode terminal 40. To configure the structure as anin-plane passive waveguide, the diffraction grating, the ohmicgate/emitter electrode layers, and any contacts to n+ and p+ regions areomitted in order to minimize waveguide loss. The waveguide ridgecross-section is formed by a combination of several mesas, which areformed by vertical/horizontal surfaces formed in the layers between thetop DBR mirror 34 and the bottom DBR mirror 12, to provide bothlaterally guiding and vertical guiding of light therein.

The heterojunction thyristor described above may be realized with amaterial system based on III-V materials (such as aGaAs/Al_(x)Ga_(1-x)As). FIG. 3A illustrates an exemplary epitaxialgrowth structure utilizing group III-V materials for realizing aheterojunction thyristor and associated optoelectrical/optical devicesin accordance with the present invention. Alternatively, strainedsilicon heterostructures employing silicon-germanium (SiGe) layers maybe used to realize the heterojunction thyristor devices and associatedoptoelectrical/optical devices described herein.

The structure of FIG. 3A can be made, for example, using known molecularbeam epitaxy (MBE) techniques. A first semiconductor layer 151 of AlAsand a second semiconductor layer 152 of GaAs are alternately deposited(with preferably at least seven pairs) upon a semi-insulating galliumarsenide substrate 149 in sequence to form the top dielectricdistributed bragg reflector (DBR) mirror 12. The number of AlAs layerswill preferably always be one greater than the number of GaAs layers sothat the first and last layers of the mirror are shown as layer 151. Inthe preferred embodiment the AlAs layers 151 are subjected to hightemperature steam oxidation to produce the compound Al_(x)O_(y) so thata mirror will be formed at the designed center wavelength. Therefore thethicknesses of layers 151 and 152 in the mirror are chosen so that thefinal optical thickness of GaAs and Al_(x)O_(y) is a quarter wavelengthof the center wavelength λ_(D). Alternatively the mirrors could be grownas alternating layers of one quarter wavelength thickness of GaAs andAlAs at the designed wavelength so that the oxidation step is not used.In that case, many more pairs are required (with typical numbers such as22 pairs) to achieve the reflectivity needed for efficient lasing.

Deposited upon the mirror is the active device structure which consistsof two HFET devices. The first of these is the above-described p-channelHFET (PHFET) 11, which has a p-type modulation doped quantum well and ispositioned with the gate terminal on the bottom (i.e. on the mirror 12just described) and the collector terminal above. The second of these isan n-channel HFET (NHFET) 13, which has an n-type modulation dopedquantum well and is positioned with the gate terminal on top and thecollector terminal below. The collector region of the NHFET device 13also functions as the collector region of the PHFET device 11. However,the collector terminal of the NHFET device 13 is a p-type contact top-type quantum well(s) disposed below (above) the collector region,while the collector terminal of the PHFET device 11 is a n-type contactto n-type quantum well(s) disposed above the collector region. Thereforea non-inverted n-channel device is stacked upon an inverted p-channeldevice to form the active device structure.

The active device layer structure begins with layer 153 of heavily N+doped GaAs of about 2000 Å thickness to enable the formation of ohmiccontacts to the gate electrode of the p-channel device. The N+ dopedGaAs layer 153 corresponds to the ohmic contact layer 14 of FIG. 2A.Deposited on layer 153 is layer 154 of n-type Al_(x1)Ga_(1-x1). As witha typical thickness of 500-3000 Å and a typical doping of 5×10¹⁷ cm⁻³.The parameter x1 is in the range between 15% and 80%, and preferably inthe range of 30%-40% for layer 154. This layer serves as part of thePHFET gate and optically as part of the lower waveguide cladding layersfor all laser, amplifier and modulator structures. Next are 4 layers(155 a, 155 b, 155 b, and 155 b) of Al_(x2)lGa_(1-x2)As. These 4 layers(collectively, 155) have a total thickness about 380-500 Å and where x2is about 15%. The first layer 155 a is about 60-80 Å thick and is dopedN+ type in the form of delta doping. The second layer 155 b is about200-300 Å thick and is undoped. The third layer 155 c is about 80 Åthick and is doped P+ type in the form of delta doping. And the fourthlayer 155 d is about 20-30 Å thick and is undoped to form a spacerlayer. This layer forms the lower separate confinement heterostructure(SCH) layer for the laser, amplifier and modulator devices. The n-typeAlGaAs layer 154 and n-type AlGaAs layer 155 a correspond to the n-typelayer(s) 16 of FIG. 2A, and the undoped AlGaAs layer 155 b correspondsto the undoped spacer layer 18 of FIG. 2A.

The next layers define the quantum well(s) that form the inversionchannel(s) during operation of the PHFET 11. For a strained quantumwell, this consists of a spacer layer 156 of undoped GaAs that is about10-25 Å thick and then combinations of a quantum well layer 157 that isabout 40-80 Å thick and a barrier layer 158 of undoped GaAs. The quantumwell layer 157 may be comprised of a range of compositions. In thepreferred embodiment, the quantum well is formed from aIn_(0.2)Ga_(0.8)AsN composition with the nitrogen content varying from0% to 5% depending upon the desired natural emission frequency. Thus,for a natural emission frequency of 0.98 μm, the nitrogen content willbe 0%; for a natural emission frequency of 1.3 μm, the nitrogen contentwill be approximately 2%; and for a natural emission frequency of 1.5μm, the nitrogen content will be approximately 4-5%. The well barriercombination will typically be repeated (for example, three times asshown), however single quantum well structures may also be used.Unstrained quantum wells are also possible. Following the last barrierof undoped GaAs is a layer 159 of undoped Al_(x2)Ga_(1-x2)As which formsthe collector of the PHFET device 11 and is about 0.5 μm in thickness.All of the layers grown thus far form the PHFET device 11 with the gatecontact on the bottom. The layers between the P+ AlGaAs layer 155 c andthe last undoped GaAs barrier layer 158 correspond to the p-typemodulation doped heterojunction QW structure 20 of FIG. 2A. UndopedAlGaAs layer 159 corresponds to the undoped spacer layer 22 of FIG. 2A.

Layer 159 also forms the collector region of the NHFET device 13.Deposited on layer 159 are two layers (collectively 160) of undoped GaAsof about 200-250 Å total thickness, which form the barrier of the firstn-type quantum well. Layer 160 is thicker than the normal barrier layerof about 100 Å because it accommodates the growth interruption to changethe growth temperature from 610° C. (as required for optical qualityAl_(x2)Ga_(1-x2)As layers) to about 530° C. for the growth of InGaAs.Therefore layer 160 includes a single layer 160 a of about 150 Å and arepeating barrier layer 160 b of about 100 Å. The next layer 161 is thequantum well of In_(0.2)Ga_(0.8)As, which is undoped and about 40-80 Åin thickness. It is noted that the n-type quantum well layer 161 neednot be of the same formulation as the p-type quantum well layer 157. Thebarrier layer 160 b of 100 Å and quantum well layer 161 may be repeated,e.g., three, times. Then there is a barrier layer 162 of about 10-30 Åof undoped GaAs which accommodates a growth interruption and a change ofgrowth temperature. Next there are four layers (collectively 163) ofAl_(x2)Ga_(1-x2)As of about 300-500 Å total thickness. These four layers(163) include a spacer layer 163 a of undoped Al_(x2)Ga_(1-x2)As that isabout 20-30 Å thick, a modulation doped layer 163 b of N+ type doping ofAl_(x2)Ga_(1-x2)As (with doping about 3.5×10¹⁸ cm⁻³) that is about 80 Åthick, a capacitor spacing layer 163 c of undoped Al_(x2)Ga_(1-x2)Asthat is about 200-300 Å thick, and a P+ type delta doped layer 163 d ofAl_(x2)Ga_(1-x2)As (with doping about 3.5×10¹⁸ cm⁻³) that is about 60-80Å to form the top plate of the capacitor. The doping species for layer163 d is preferably carbon (C) to ensure diffusive stability. Incontrast to layer 163 b which is always depleted, layer 163 d shouldnever be totally depleted in operation. Layers 163 d and 163 b form thetwo plates of a parallel plate capacitor which forms the field-effectinput to all active devices. For the optoelectronic device operation,layer 163 is the upper SCH region. Layer 163 must be thin to enable veryhigh frequency operation. In the illustrated embodiment, for atransistor cutoff frequency of 40 GHz, a thickness of 300 Å would beused, and for 90 GHz a thickness of 200 Å would be more appropriate. Thelayers between the undoped GaAs barrier layer 160 a and the N+ AlGaAslayer 163 b correspond to the n-type modulation doped heterojunction QWstructure 24 of FIG. 2A. Undoped AlGaAs layer 163 c corresponds to theundoped spacer layer 26 of FIG. 2A.

One or more layers (collectively 164) of p-type Al_(x1)Ga_(1-x1)As aredeposited next to form part of the upper waveguide cladding layer forthe laser, amplifier and modulator devices. It has a typical thicknessof 500-1500 Å. Layer 164 may have a first thin sublayer 164 a of, e.g.,10-20 Å thickness and having a P+ typical doping of 10¹⁹ cm⁻³. A secondsublayer 164 b has a P doping of 1×10¹⁷−5×10¹⁷ cm⁻³ and a typicalthickness of 700 Å. The parameter X1 of layer 164 is preferably about70%. The p-type layers 163 b, 164 a, 164 b correspond to the p-typelayer(s) 28 of FIG. 2A.

Deposited next is an ohmic contact layer 165 (which may comprise asingle layer of GaAs or a combination of GaAs (165 a) and InGaAs (165 b)as shown), which is about 50-100 Å thick and doped to a very high levelof P+ type doping (about 1×10²⁰ cm⁻³) to enable the best possible ohmiccontact.

The band diagram of the FIG. 3A structure is shown in FIG. 3B.

To form a resonant cavity device where light is input into and emittedfrom the device laterally (i.e., from a direction normal to the crosssection of FIG. 3A), a diffraction grating (as described in more detailin U.S. Pat. No. 6,021,243, incorporated by reference above in itsentirety) and top DBR mirror are formed over the active device structuredescribed above. When the heterojunction thyristor device is operatingin the lasing mode, the diffraction grating performs the function ofdiffracting light produced by the vertical cavity into light propagatinglaterally in a waveguide which has the top DBR mirror and bottom DBRmirror as waveguide cladding layers and which has lateral confinementregions (typically formed by implants as described herein in moredetail). When the heterojunction thyristor device is operating in theoptical detection mode, the diffraction grating performs the function ofdiffracting incident light that is propagating in the lateral directioninto the vertical cavity mode, where it is absorbed resonantly in thevertical cavity.

Alternatively, light may enter and exit the resonant vertical cavityvertically through an optical aperture in the top surface of the device.In this case, the diffraction grating is omitted, the top DBR mirrordefines a cavity for the vertical emission and absorption of light, andthe device operates as a vertical cavity surface emittinglaser/detector. The distance between the top DBR mirror and bottom DBRmirror preferably represents an integral number of ¼ wavelengths at thedesignated wavelength. Preferably, the thickness of layer 164 or 159 isadjusted to enable this condition.

Using the structure described above with respect to FIGS. 3A and 3B, aheterojunction thyristor can be realized as shown in FIG. 3C. To connectto the anode of the device, alignment marks (not shown) are defined byetching, and then a layer of Si₃N₄ or Al₂O₃ or other suitable dielectric(not shown) is deposited to act as protection for the surface layer andas a blocking layer for subsequent ion implants. This dielectric layeralso forms the first layer of the top DBR mirror. Then an ion implant175 of n-type is performed using a photomask that is aligned to thealignments marks, and an optical aperture is defined by the separationbetween the implants 175. The implants 175 create a p-n junction in thelayers between the n-type quantum wells and the surface, and theaperture between the implants defines the region in which the currentmay flow, and therefore the optically active region 177 as shown. Thecurrent cannot flow into the n-type implanted regions 175 because of thebarrier to current injection. The current flow trajectory is shown inFIG. 3C as arrows. The laser threshold condition is reached before thevoltage for turn-on of this barrier. Following the implant, therefractory anode terminals 36A and 36B (which collectively form theanode terminal 36 of the device) are deposited and defined.

N+ ion implants 170 are used to form self-aligned channel contacts tothe n-type QW inversion channel(s). More specifically, the N+ implantsare used as an etch stop to form a mesa via etching down (for example,to layer 163 c) near the n-type QW channel(s). The N+ ion implants 170are electrically coupled to the injector terminals 38A and 38B (whichcollectively form the injector terminal 38 of the device). The injectorterminals 38A and 38B are preferably formed via deposition of an n-typeAu alloy metal on the N+ ion implants 170 to form ohmic contactsthereto. In the event that injector terminals of the device are coupledto the p-type QW inversion channel(s), P+ ion implants (not shown) areused to form self-aligned channel contacts to the p-type QW inversionchannel(s). In this case, injector terminals 38A and 38B are preferablyformed via deposition of an p-type Au alloy metal on the P+ ion implantsto form ohmic contacts thereto.

Alternatively, first injector terminals may be operably coupled to then-type QW channel(s) while second injector terminals are operablycoupled to the P-type QW channel(s). These channel contacts enableswitching of the thyristor with n-type and/or p-type high impedancesignals via the injector terminals. Connection to the cathode terminals40A and 40B (which collectively form the cathode terminal 40 of thedevice) is provided by etching to the N+ bottom layer 153, anddepositing a metal layer (for example AuGe/Ni/Au) to form an ohmiccontact to N+ bottom layer 153. The resulting structured is isolatedfrom other devices by etching down to the substrate 149. The structureis then subject to rapid thermal anneal (RTA) to activate the implants.

To form a device suitable for in-plane optical injection into a resonantvertical cavity and/or in-plane optical emission from the resonantvertical cavity, a diffraction grating (as described in more detail inU.S. Pat. No. 6,021,243, incorporated by reference above in itsentirety) and top DBR mirror are formed on this structure as describedabove. To form a device suitable for vertical optical injection into(and/or optical emission from) a resonant vertical cavity, thediffraction grating is omitted. The diffraction grating, when used, iscreated over the active device structure described above. The top DBRmirror is preferably created by the deposition of one or more dielectriclayer pairs (179,180), which typically comprise SiO₂ and a highrefractive index material such as GaAs, Si, or GaN, respectively.

FIG. 3D illustrates an alternate epitaxial growth structure utilizinggroup III-V materials for realizing a heterojunction thyristor andassociated optoelectrical/optical devices in accordance with the presentinvention. The structure of FIG. 3D can be made, for example, usingknown molecular beam epitaxy (MBE) techniques. Similar to the growthstructure of FIG. 3A, a first semiconductor layer 151 of AlAs and asecond semiconductor layer 152 of GaAs are alternately deposited (withpreferably at least seven pairs) upon a semi-insulating gallium arsenidesubstrate 149 in sequence to form the top dielectric distributed braggreflector (DBR) mirror 12. The number of AlAs layers will preferablyalways be one greater than the number of GaAs layers so that the firstand last layers of the mirror are shown as layer 151. In the preferredembodiment the AlAs layers 151 are subjected to high temperature steamoxidation to produce the compound Al_(x)O_(y) so that a mirror will beformed at the designed center wavelength. Therefore the thicknesses oflayers 151 and 152 in the mirror are chosen so that the final opticalthickness of GaAs and Al_(x)O_(y) is a quarter wavelength of the centerwavelength λ_(D). Alternatively the mirrors could be grown asalternating layers of one quarter wavelength thickness of GaAs and AlAsat the designed wavelength so that the oxidation step is not used. Inthat case, many more pairs are required (with typical numbers such as 22pairs) to achieve the reflectivity needed for efficient lasing.

Deposited upon the mirror is the active device structure which consistsof two HFET devices. The first of these is the above-described p-channelHFET (PHFET) 11, which has one or more p-type modulation doped quantumwells and is positioned with the gate terminal on the bottom (i.e. onthe mirror 12 just described) and the collector terminal above. Thesecond of these is an n-channel HFET (NHFET) 13, which has one or moren-type modulation doped quantum wells and is positioned with the gateterminal on top and the collector terminal below. The collector regionof the NHFET device 13 also functions as the collector region of thePHFET device 11. However, the collector terminal of the NHFET device 13is a p-type contact to p-type quantum well(s) disposed below (above) thecollector region, while the collector terminal of the PHFET device 11 isa n-type contact to n-type quantum well(s) disposed above the collectorregion. Therefore a non-inverted n-channel device is stacked upon aninverted p-channel device to form the active device structure.

The active-device layer structure begins with layer 153 of N+ type GaAsthat enables the formation of ohmic contacts thereto (for example, whencontacting to the cathode terminal of a heterojunction thyristor device,the gate terminal of an inverted p-channel HFET device, thesub-collector terminal of an n-channel HFET device, or the emitterterminal of a p-type quantum-well-base bipolar transistor device). Layer153 has a typical thickness of 1000-2000 Å and a typical n-type dopingof 3.5×10¹⁸ cm⁻³. The N+ doped GaAs layer 153 corresponds to the ohmiccontact layer 14 of FIG. 2A. Deposited on layer 153 is layer 166 a ofn-type AlAs having a typical thickness of 30-200 Å and a typical n-typedoping of 3.5×10¹⁸ cm⁻³. One constraint upon the thickness and thedoping of this layer 166A is that it should not be depleted in any rangeof operation of the device, i.e. the total doping in this layer shouldexceed the total doping charge contained in the layer 155 c describedbelow. This layer 166 a serves optically as the lower waveguide claddinglayers for all laser, amplifier and modulator structures. In addition,it also acts as a etch stop layer (described below in more detail) whenforming contacts to the ohmic contact layer 153. Another constraint onthe thickness of layer 166 a is that it must be made sufficiently thinto enable hole current to flow through it by tunneling. In this manner,the thickness of this layer 166 a determines the current gain of aninverted p-type quantum-well-base bipolar transistor device realized inthis growth structure. Next is a layer 166 b of undoped GaAs having atypical thickness of 6-20 Å. This layer 166 b serves to preventoxidation of the layer 166 a during subsequent oxidation operations(e.g., where the bottom DBR mirror layers 151/152 are oxidized). Inaddition, undoped GaAs layer 166 b is advantageous in a single aluminumeffusion cell MBE system because it accommodates a growth interruptionto change the growth temperature between layers 166 a and 155 b asrequired.

Next are three layers (155 b, 155 c, and 155 d) of Al_(x2)Ga_(1-x2)As.These three layers have a total thickness about 300-500 Å and where x2is about 15%. The first layer 155 b is about 200-300 Å thick and isundoped. The second layer 155 c is about 80 Å thick and is doped P+ typein the form of delta doping with a typical concentration of 3.5×10¹⁸cm⁻³. And the third layer 155 d is about 20-30 Å thick and is undoped.This layer 155 d forms the lower separate confinement heterostructure(SCH) layer for the laser, amplifier and modulator devices. The N+ AlAslayer 166 a corresponds to the n-type layer 16 of FIG. 2A, and theundoped GaAs layer 166 b and the undoped GaAs layer 155 b corresponds tothe undoped spacer layer 18 of FIG. 2A. To realize a p-typequantum-well-base bipolar transistor device (and/or a p-channel HFET)with a cutoff frequency of about 40 GHz, the thickness of layers 166 band 155 b are preferably on the order of 300 Å. And to realize a p-typequantum-well-base bipolar transistor (and/or a p-channel HFET) with acutoff frequency of about 90 GHz, the thickness of layers 166 b and 155b are preferably on the order of 250 Å.

The next layers define the quantum well(s) that form the inversionchannel(s) during operation of the PHFET 11. For a strained quantumwell, this consists of a spacer layer 156 of undoped GaAs that is about10-25 Å thick and then combinations of a quantum well layer 157 (that isabout 40-80 Å thick) and a barrier layer 158 of undoped GaAs. Thequantum well layer 157 may be comprised of a range of compositions. Inthe preferred embodiment, the quantum well is formed from aIn_(0.2)Ga_(0.8)AsN composition with the nitrogen content varying from0% to 5% depending upon the desired natural emission frequency. Thus,for a natural emission frequency of 0.98 μm, the nitrogen content willbe 0%; for a natural emission frequency of 1.3 μm, the nitrogen contentwill be approximately 2%; and for a natural emission frequency of 1.5μm, the nitrogen content will be approximately 4-5%. The well-barriercombination will typically be repeated (for example, three times asshown) to define the quantum wells that form the inversion channelsduring operation of the PHFET 11 (however single quantum well structuresare also possible). Unstrained quantum wells are also possible.Following the last barrier of undoped GaAs is a layer 167 of undopedGaAs and a layer 159 of undoped Al₂Ga_(1-x2)As. The undoped GaAs layer167 has a typical thickness of 250-500 Å, and the undoped Al₂Ga_(1-x2)Aslayer 159 has a typical thickness of 0.51 μm. These layers 167 and 159form the collector of the PHFET device 11. The purpose of the GaAs layer167 is to accommodate a change in the growth temperature from about 530°C. (as required for the InGaAs quantum well structure of layer 157) toabout 610° C. (as required for Al_(x2)Ga_(1-x2)As layer 159). Layer 167performs no electrical purpose and so it should be electrically totallytransparent to all current flows. Therefore, layer 167 is thin enoughthat currents may conduct through it by tunneling with negligiblevoltage drop. All of the layers grown thus far form the PHFET device 11with the gate contact on the bottom. The layers between the P+ AlGaAslayer 155 c and the last undoped GaAs barrier layer 158 correspond tothe p-type modulation doped heterojunction QW structure 20 of FIG. 2A.Undoped GaAs layer 167 and undoped AlGaAs layer 159 corresponds to theundoped spacer layer 22 of FIG. 2A.

Layers 167 and 159 also form the collector region of the NHFET device13. Deposited on layer 159 are two layers 160 a, 160 b (collectively160) of undoped GaAs of about 200-250 Å total thickness, which form thebarrier of the first n-type quantum well. Layer 160 is thicker than thenormal barrier layer of about 100 Å because it accommodates a change ofthe growth temperature from 610° C. (as required for the Al₂Ga_(1-x2)Aslayer 159) to about 530° C. (as required for the In_(0.2)Ga_(0.8)Asquantum well layer 161). The next layer 161 is the quantum well ofIn_(0.2)Ga_(0.8)As, which is undoped and about 40-80 Å in thickness. Thequantum well layer 161 may be comprised of a range of compositions asdescribed above with respect to the quantum well layer 157. In thepreferred embodiment, the quantum well is formed from anIn_(0.2)Ga_(0.8)AsN composition with the nitrogen content varying from0% to 5% depending upon the desired natural emission frequency. It isnoted that the n-type quantum well layer 161 need not be of the sameformulation as the p-type quantum well layer 157. The barrier-wellcombination will typically be repeated (for example, three times asshown) to define the quantum wells that form the inversion channel(s)during operation of the NHFET 13. Then there is a barrier layer 162 ofabout 10-30 Å of undoped GaAs which accommodates a growth interruptionand a change of growth temperature.

Next there are three layers (163 a, 163 b, 163 c) of Al_(x2)Ga_(1-x2)Asof about 300-500 Å total thickness. These three layers include a spacerlayer 163 a of undoped Al_(x2)Ga_(1-x2)As that is about 20-30 Å thick, amodulation doped layer 163 b of N+ type doping of Al_(x2)Ga_(1-x2)As(with doping about 3.5×10¹⁸ cm⁻³) that is about 80 Å thick, and a spacerlayer 163 c of undoped Al_(x2)Ga_(1-x2)As that is about 200-300 Å thick.Next is a layer 168 a of undoped GaAs that is about 6-20 Å thick, and aP+ type doped layer 168 b of AlAs (with doping about 3.5×10¹⁸ cm⁻³) thatis about 300 Å. In contrast to layer 163 b which is always depleted,layer 168 b should never be totally depleted in operation (i.e., thetotal doped charge in layer 168 b should always exceed that in layer 163b). Layers 168 b and 163 b (and the undoped spacer layers 163 c and 168a therebetween) form the two plates of a parallel plate capacitor whichforms the field-effect input to all active devices. For theoptoelectronic device operation, layer 163 a is the upper SCH region.Layer 168 b also acts as a etch stop layer (described below in moredetail) when forming contacts to the N-type inversion channel(s) of theNHFET 13 (for example, when contacting to the N-channel injectorterminal(s) of a heterojunction thyristor device, the source/drainterminals of an n-channel HFET device, the base terminal of an n-typep-type quantum-well-base bipolar transistor, or the collector terminalof a p-type quantum-well-base bipolar transistor device). Layer 168 aserves to prevent oxidation of previous layers 163 a, 163 b, 163 c ofAl_(x2)Ga_(1-x2)As during subsequent oxidation operations (e.g., wherethe bottom DBR mirror layers are oxidized). Moreover, similar to layer166 b, layer 168 a must be made sufficiently thin to enable electroncurrent to flow through it by tunneling. In this manner, the thicknessof this layer 168 a determines the current gain of an n-typequantum-well-base bipolar transistor device realized in this growthstructure. In addition, undoped GaAs layer 168 a is advantageous in asingle aluminum effusion cell MBE system because it accommodates agrowth interruption to change the growth temperature between layers 163c and 168 b as required. The layers between the undoped GaAs barrierlayer 160 a and the N+ AlGaAs layer 163 b correspond to the n-typemodulation doped heterojunction QW structure 24 of FIG. 2A. UndopedAlGaAs layer 163 c and undoped GaAs layer 168 a corresponds to theundoped spacer layer 26 of FIG. 2A. To realize an n-typequantum-well-base bipolar transistor (and/or an n-channel HFET) with acutoff frequency of about 40 GHz, the thickness of layers 163 c and 168a are preferably on the order of 300 Å. And to realize an n-typequantum-well-base bipolar transistor (and/or an n-channel HFET) with acutoff frequency of about 90 GHz, the thickness of layers 163 c and 168a are preferably on the order of 250 Å.

A layer 164 of p-type GaAs is deposited next to form part of the upperwaveguide cladding layer for the laser, amplifier and modulator devices.It also forms a spacer layer in which to accommodate the apertureimplant which steers the current into the VCSEL active region. It shouldprovide a low resistance access to the top contact. It has a typicalthickness of 300 Å. The p-type layers 168 b and 164 correspond to thep-type layer(s) 28 of FIG. 2A.

Deposited next is an ohmic contact layer 165 (which may comprise asingle layer of GaAs or a combination of GaAs (165 a) and InGaAs (165 b)as shown). In the illustrative embodiment shown, GaAs layer 165 a isabout 50-100 Å thick and doped to a very high level of P+ type doping(about 1×10²⁰ cm⁻³) and InGaAs layer 165 b is about 25-50 Å thick anddoped t very high level of P+ type doping (about 1×10²⁰ cm⁻³) to enablethe best possible ohmic contact.

The band diagram of the FIG. 3D structure is shown in FIG. 3E.

To form a resonant cavity device where light is input into and emittedfrom the device laterally (i.e., from a direction normal to the crosssection of FIG. 3D), a diffraction grating (as described in more detailin U.S. Pat. No. 6,021,243, incorporated by reference above in itsentirety) and top DBR mirror are formed over the active device structuredescribed above. When the heterojunction thyristor device is operatingin the lasing mode, the diffraction grating performs the function ofdiffracting light produced by the vertical cavity into light propagatinglaterally in a waveguide which has the top DBR mirror and bottom DBRmirror as waveguide cladding layers and which has lateral confinementregions (typically formed by implants as described herein in moredetail). When the heterojunction thyristor device is operating in theoptical detection mode, the diffraction grating performs the function ofdiffracting incident light that is propagating in the lateral directioninto the vertical cavity mode, where it is absorbed resonantly in thevertical cavity.

Alternatively, light may enter and exit the resonant vertical cavityvertically through an optical aperture in the top surface of the device.In this case, the diffraction grating is omitted, the top DBR mirrordefines a cavity for the vertical emission and absorption of light, andthe device operates as a vertical cavity surface emittinglaser/detector. The distance between the top DBR mirror and bottom DBRmirror preferably represents an integral number of ¼ wavelengths at thedesignated wavelength. Preferably, the thickness of layer 164 or 159 isadjusted to enable this condition.

Using the structure described above with respect to FIGS. 3D and 3E, aheterojunction thyristor can be realized as shown in FIG. 3F. To connectto the anode of the device, alignment marks (not shown) are defined byetching, and then a layer of Si₃N₄ or Al₂O₃ or other suitable dielectric(not shown) is deposited to act as protection for the surface layer andas a blocking layer for subsequent ion implants. Preferably, thisdielectric layer also forms the first layer of the top DBR mirror. Thenan ion implant 175 of n-type is performed using a photomask that isaligned to the alignments marks, and an optical aperture is defined bythe separation between the implants 175. The implants 175 create a p-njunction in the layers between the n-type quantum well(s) and thesurface, and the aperture between the implants defines the region inwhich the current may flow, and therefore the optically active region177 as shown. The current cannot flow into the n-type implanted regions175 because of the barrier to current injection. The current flowtrajectory is shown in FIG. 3F as arrows. The laser threshold conditionis reached before the voltage for turn-on of this barrier. Following theimplant 175, the refractory anode terminals 36A and 36B (whichcollectively form the anode terminal 36 of the device) are deposited anddefined.

Then an ion implant 170 of n+-type is performed using a photomask thatis aligned to the alignments marks, to thereby form contacts to then-type QW inversion channel(s). During this operation, a chlorine-basedgas mixture that includes fluorine is used as an etchant to etch down tothe etch-stop layer 168 b. The etch rate through the InGaAs layer 165 band GaAs layers (165 a and 164) is fairly rapid. However, because of thepresence of fluorine in the etchant, the etch rate decreases drasticallywhen the AlAs layer 168 b is encountered. This is because the AlAs layer168 b has a high percentage of Aluminum, which forms AIF in the presenceof the etch mixture. The AIF deposits on the surface of the structureand prevents further etching (because it is non-volatile and not etchedby any of the conventional etchants). In this manner, the AlAs layer 168b operates as an etch stop layer. This layer is then easily dissolved inde-ionized (DI) water or wet buffered hydrofluoric acid (BHF) to formmesas at the undoped GaAs layer 168 a. The resulting mesas at theundoped GaAs layer 168 a is subject to the N+ ion implants 170, whichare electrically coupled to the N-channel injector terminals 38A and38B. The N-channel injector terminals 38A and 38B are preferably formedvia deposition of an n-type Au alloy metal on the N+ ion implants 170 toform ohmic contacts thereto.

Then an ion implant 171 of p+-type is performed using a photomask thatis aligned to the alignments marks, to thereby form contacts to thep-type QW inversion channel(s). During this operation, mesas are formedby etching preferably down to the undoped GaAs layer 158. The resultingmesas are then subject to P+ ion implants 171, which are electricallycoupled to the P-channel injector terminals 38C and 38D. The P-channelinjector terminals 38C and 38D are preferably formed via deposition ofan p-type alloy metal on the P+ ion implants 171 to form ohmic contactsthereto.

In alternative embodiments, the P+ ion implants 171 (and correspondingP-channel injector terminals 38C and 38D) may be omitted. In such aconfiguration, the N-channel injector terminals 38A and 38B (which arecoupled to the n-type inversion QW channel(s) of the NHFET device by theN+ ion implants 170) are used to control charge in such n-type inversionQW channel(s) as described herein. In yet another alternativeembodiment, the N+ ion implants 170 (and corresponding N-channelinjector terminals 38A and 38B) may be omitted. In such a configuration,the P-channel injector terminals 38C and 38D (which are coupled to thep-type inversion QW channel(s) of the PHFET 11 device by the P+ ionimplants 171) are used to control charge in such p-type inversion QWchannel(s) as described herein.

Connection to the cathode terminals 40A and 40B the device is made byetching with a chlorine-based gas mixture that includes fluorine. Thisetch is performed down to the AlAs etch stop layer 166 a as describedabove. This layer 166 a is then easily dissolved in de-ionized (DI)water or wet buffered hydrofluoric acid (BHF) to form resulting mesas inthe N+ layer 153. A metal layer (for example AuGe/Ni/Au) is deposited onthe mesas at the N+ layer 153 to formed an ohmic contact thereto. Theresulting structured is isolated from other devices by etching down tothe substrate 149. The structure is then subject to rapid thermal anneal(RTA) to activate the implants.

To form a device suitable for in-plane optical injection into a resonantvertical cavity and/or in-plane optical emission from the resonantvertical cavity, a diffraction grating 32 (as described in more detailin U.S. Pat. No. 6,021,243, incorporated by reference above in itsentirety) and top DBR mirror is deposited on this structure as describedabove. To form a device suitable for vertical optical injection into(and/or optical emission from) a resonant vertical cavity, thediffraction grating 32 is omitted. The top DBR mirror is preferablycreated by the deposition of one or more dielectric layer pairs(179,180), which typically comprise SiO₂ and a high refractive indexmaterial such as GaAs, Si, or GaN, respectively.

Turning now to FIGS. 4A through 5F, a heterojunction thyristor device,which is configured to convert a digital optical signal to acorresponding digital electrical signal as described above, is used asthe basis for a photonic digital-to-analog converter that converts adigital word encoded by a plurality of input digital optical signals toan output analog electrical signal corresponding to the digital word.

A first exemplary embodiment of the photonic digital-to analog converter201 is shown in FIG. 4A. The input digital optical signals synchronouslyencode a plurality of bits of information that are logically arrangedfrom a most-significant-bit (MSB) to a least-significant-bit (LSB) asshown in FIGS. 4A and 4B(i). These bits form a digital word. In theexemplary embodiment shown, four (4) input digital optical signalsencode four (4) bits of information logically arranged from amost-significant-bit (MSB) to a least-significant-bit (MSB₋₃). Aplurality of heterojunction thyristor devices (4 shown as 202 _(MSB),202 _(MSB-1), 202 _(MSB-2), 202 _(MSB-3)), each corresponding to adifferent input digital optical signal/bit, are formed in resonantcavities on at least one substrate. Preferably, the pluralityheterojunction thyristor devices are integrally formed in resonantcavities on a common substrate. In addition, a plurality of voltagedivider networks (4 shown as 204 _(MSB), 204 _(MSB-1), 204 _(MSB-2), 204_(MSB-3)), each corresponding to a different heterojunction thyristordevice and input digital optical signal/bit, are operably coupled to thecathode terminal of the heterojunction thyristor devices (202 _(MSB),202 _(MSB-1), 202 _(MSB-2), 202 _(MSB-3)). A summing circuit 206 isoperably coupled to the voltage divider networks (204 _(MSB), 204_(MSB-1), 204 _(MSB-2), 204 _(MSB-3)) as shown. A timing signalgenerator 207 generates an electrical timing signal A and an electricaltiming signal B as shown in FIG. 4B(ii) and 4B(iii), respectively. Theelectrical timing signal A includes a sampling clock pulse whoseduration defines a sampling period that overlaps the bits of informationencoded in the input digital optical signals. The electrical timingsignal B includes a summing clock pulse whose duration defines a summingperiod that is subsequent to the sampling period.

During conversion operations, the plurality of input optical signals areinjected into the resonant cavities, where such signals are resonantlyabsorbed by the heterojunction thyristor devices (204 _(MSB), 202_(MSB-1), 202 _(MSB-2), 202 _(MSB-3)). Each heterojunction thyristordevice is configured (as described above with respect FIGS. 2B1, 2B3,2C, and 2D1) to operate during a given sampling period to convert theinput digital optical signal supplied thereto to a corresponding digitalelectrical signal. The voltage divider networks (204 _(MSB), 204_(MSB-1), 204 _(MSB-2), 204 _(MSB-3)) are configured to scale themagnitude of the digital electrical signals produced by theheterojunction thyristor devices (202 _(MSB), 202 _(MSB-1), 202_(MSB-2), 202 _(MSB-3)) to produce output electrical signals (V_(MSB),V_(MSB-1), V_(MSB-2), V_(MSB-3)) whose magnitude corresponds to the bitsencoded by the input digital optical signals. For example, the voltagedivider networks (204 _(MSB), 204 _(MSB-1), 204 _(MSB-2), 204 _(MSB-3))shown in FIG. 4A scale the magnitude of the digital electrical signalsproduced by the heterojunction thyristor devices (202 _(MSB), 202_(MSB-1), 202 _(MSB-2), 202 _(MSB-3)) by factors of 1, ½, ¼, and ⅛,respectively, to produce output electrical signals as shown in thefollowing table I.

TABLE I Output Voltage Level when the Voltage Level when the Electricalcorresponding optical bit is corresponding optical bit is Signal ON OFFV_(MSB) V_(D) 0 V_(MSB-1) V_(D)/2 0 V_(MSB-2) V_(D)/4 0 V_(MSB-3)V_(D)/8 0The scaling function for a given voltage divider network can berepresented generally by a scale factor of (½^(X)), where X is thedifference between the position of the corresponding bit and the MSB.

The summing circuitry 206 operates during the summing period to sum theoutput electrical signals (V_(MSB), V_(MSB-1), V_(MSB-2), V_(MSB-3))produced by the voltage divider networks (204 _(MSB), 204 _(MSB-1), 204_(MSB-2), 204 _(MSB-3)). The result of the summing operation performedby the summing circuitry 205, which is output from the summing circuitry206, is an analog electric signal whose magnitude corresponds to thedigital word encoded by the bits of the input digital optical signals.For example, the summing circuit 206 of FIG. 4A sums the four (4) outputelectrical signals to produce a resultant analog signal whose magnitudecorresponds to the digital word encoded by the four (4) bits of theinput digital optical signals as shown in the following table II.

TABLE II Analog Output MSB MSB₋₁ MSB₋₂ MSB₋₃ (V) (Optical) (Optical)(Optical) (Optical) (Elect) 0 (OFF) 0 0 O 0.0 0 0 0 1 (ON)  V_(D)/8 0 01 0  V_(D)/4 0 0 1 1  3V_(D)/8 0 1 0 0  V_(D)/2 0 1 0 1  5V_(D)/8 0 1 10  3V_(D)/4 0 1 1 1  7V_(D)/8 1 0 0 0  V_(D) 1 0 0 1  9V_(D)/8 1 0 1 0 5V_(D)/4 1 0 1 1 11V_(D)/8 1 1 0 0  3V_(D)/2 1 1 0 1 13V_(D)/8 1 1 1 0 7V_(D)/4 1 1 1 1 15V_(D)/8

Preferably, the summing circuitry 206 includes a chain of two-portadding nodes 208 and sample/hold circuits 210 arranged as pairs, eachpair corresponding to a different voltage divider network (204 _(MSB-1),204 _(MSB-2), 204 _(MSB-3)) as shown. In this configuration, the outputelectrical signal generated by a given voltage divider network issupplied to an input node of the corresponding two-port adding node 208.In addition, a single sample/hold circuit 211 supplies an outputelectrical signal (V_(MSB)) whose magnitude corresponds to the MSB tothe chain of two-port adding nodes 208 and sample/hold circuits 210. Thesample and hold circuits 210 of this chain and the single sample andhold circuit 211 are activated during the summing period (which isdefined by the duration of the summing clock pulse in the timing signalB as shown in FIG. 4B(iii)) to thereby effectuate the summing operation,which produces the output analog electrical signal as described above.In this configuration, each sample and hold circuit 210 and 211 includesan input capacitance that operates to store the electrical signalgenerated by a given voltage divider network and supplied thereto duringa given sampling period for summation during the subsequent summingperiod.

Preferably, such conversion operations are repeated for another digitalword that is subsequently encoded by the input digital optical signalssupplied to the converter 201 as described above.

In the embodiment of FIG. 4A, each heterojunction thyristor deviceoperates to convert the input digital optical signal supplied thereto toa corresponding digital electrical signal as described above withrespect to FIGS. 2B1, 2B3, 2C, and 2D1. In this embodiment, the lengthand width of the given heterojunction thyristor device are sized suchthat it switches from a non-conducting/OFF state to a conducting/ONstate when the combination of i) the injection of electrical energy intothe QW channel(s) of the device from the electrical sampling clock pulseand ii) absorption of optical energy in the QW channel(s) of the devicefrom the ON pulse of the input optical signal produces a channel currentthat exceeds the bias current I_(BIAS) such that charge in the QWchannel(s) of the device build to a level that is greater than thecritical switching charge Q_(CR). When the electrical sampling clockpulse terminates, the bias current I_(BIAS) reduces the charge in QWchannel(s) of the device to a level below the holding charge Q_(H),thereby causing the device to switch from the conducting/ON state to thenon-conducting/OFF state. Note that the device does not switch from thenon-conducting/OFF state to the conducting/ON state in the event thateither the electrical sampling clock pulse is not present or the inputoptical signal represents the OFF logic level. This occurs because thesesignals alone are not sufficient to produce the critical switchingcharge Q_(CR).

In another aspect of the present invention, the heterojunction thyristordevice can be configured to operate as an electrically-controlledsampling device (e.g., electrically-controlled switch), which issuitable for use in a sample and hold circuit (such as sample and holdcircuits 210/211 in FIG. 4A) and in many other signal processingapplications (such as switched capacitance filters and switchedcapacitance waveform generators). Two possible configurations are shownin FIGS. 4C and 4D, respectively.

In FIG. 4C, a first p-channel injector terminal 222 (the electricalinput terminal) and a second p-channel injector terminal 224 (theelectrical output terminal) are operably coupled to opposite ends of thep-type QW channel(s) of structure 20. An electrical sampling clock pulsein the form of a downward running electrical pulse is supplied to then-channel injector terminal(s) 226 of the device. In addition, a biascurrent source is coupled to the n-channel injector terminal(s) 226 anddraws charge from the n-type QW channel(s) to the positive supplyvoltage potential V_(D). The anode terminal 228 is forward biased (e.g.biased positively) with respect to the cathode terminal 230. The lengthand width of the device are sized such that it switches from anon-conducting/OFF state to a conducting/ON state when the electricalenergy injected into the n-type QW channel(s) of structure 24 by theelectrical sampling clock pulse produces a channel current that exceedsthe bias current I_(BIAS) such that charge in the n-type QW channel(s)of structure 24 builds to a level that is greater than the criticalswitching charge Q_(CR). When the electrical sampling clock pulseterminates, the bias current I_(BIAS) reduces the charge in the n-typeQW channel(s) to a level below the holding charge Q_(H), thereby causingthe device to switch from the conducting/ON state to thenon-conducting/OFF state. Note that the device does not switch from thenon-conducting/OFF state to the conducting/ON state in the event thatthe electrical sampling clock pulse is not present. This occurs becausethere is no injection of electrical energy into the QW channel(s) of thedevice to produce the critical switching charge Q_(CR).

When the device is operating in the non-conducting/OFF state, theelectrical input terminal 222 is electrically isolated from theelectrical output terminal 224. However, when the device is operating inthe conducting/ON state, the electrical input terminal 222 iselectrically coupled to the electrical output terminal 224 (and there isminimal potential voltage differences between input terminal 222 andoutput terminal 224). In this manner, the heterojunction thyristordevice operates as a sampling device (e.g., switch) that is selectivelyactivated and deactivated by an electric control signal (e.g., thesample clock pulse).

In FIG. 4D, a first n-channel injector terminal 232 (the electricalinput terminal) and a second n-channel injector terminal 234 (theelectrical output terminal) are operably coupled to opposite ends of then-type QW channel(s) of structure 24. An electrical sampling clock pulsein the form of a upward running electrical pulse is supplied to thep-channel injector terminal(s) 236 of the device. In addition, a biascurrent source is coupled to the p-channel injector terminal(s) 236 anddraws charge from the p-type QW channel(s) to the ground potential. Theanode terminal 238 is forward biased (e.g. biased positively) withrespect to the cathode terminal 240. The length and width of the deviceare sized such that it switches from a non-conducting/OFF state to aconducting/ON state when the electrical energy injected into the p-typeQW channel(s) of structure 20 by the electrical sampling clock pulseproduces a channel current that exceeds the bias current I_(BIAS) suchthat charge in the p-type QW channel(s) of structure 20 builds to alevel that is greater than the critical switching charge Q_(CR). Whenthe electrical sampling clock pulse terminates, the bias currentI_(BIAS) reduces the charge in the p-type QW channel(s) to a level belowthe holding charge Q_(H), thereby causing the device to switch from theconducting/ON state to the non-conducting/OFF state. Note that thedevice does not switch from the non-conducting/OFF state to theconducting/ON state in the event that the electrical sampling clockpulse is not present. This occurs because there is no injection ofelectrical energy into the QW channel(s) of the device to produce thecritical switching charge Q_(CR).

When the device is operating in the non-conducting/OFF state, theelectrical input terminal 232 is electrically isolated from theelectrical output terminal 234. However, when the device is operating inthe conducting/ON state, the electrical input terminal 232 iselectrically coupled to the electrical output terminal 234 (and there isminimal potential voltage differences between input terminal 232 andoutput terminal 234). In this manner, the heterojunction thyristordevice operates as a sampling device (e.g., switch) that is selectivelyactivated and deactivated by an electric control signal (e.g., thesample clock pulse).

Another photonic digital-to analog converter 201′ is shown in FIG. 4E.Similar to the embodiment of FIG. 4A, the input digital optical signalssynchronously encode a plurality of bits of information that arelogically arranged from a most-significant-bit (MSB) to aleast-significant-bit (LSB) as shown in FIG. 4B(i). These bits form adigital word. In the exemplary embodiment shown, four (4) input digitaloptical signals encode four (4) bits of information logically arrangedfrom a most-significant-bit (MSB) to a least-significant-bit (MSB₋₃). Aplurality of heterojunction thyristor devices (4 shown as 302 _(MSB),302 _(MSB-1), 302 _(MSB-2), 302 _(MSB-3)), each corresponding to adifferent input digital optical signal/bit, are formed in resonantcavities on at least one substrate. Preferably, the plurality ofheterojunction thyristor devices are integrally formed in resonantcavities on a common substrate. Each heterojunction thyristor device isconfigured as a sampling device (e.g., switch) in a manner similar tothe sampling device described below with respect to FIGS. 6A and 6B,whereby the n-channel injector terminals form the electrical inputterminal 304 and the electrical output terminal 306. A bias currentsource is coupled to the p-channel injector terminal(s) 308 and drawscharge from the QW channel(s) coupled thereto. The anode terminal 310 isforward biased (e.g. biased positively) with respect to the cathodeterminal 312. As described below in detail with FIGS. 6A and 6B, thelength and width of each heterojunction-thyristor-based sampling deviceare sized such that it switches from a non-conducting/OFF state to aconducting/ON state when the combination of i) the injection ofelectrical energy into the QW channel(s) of the device from theelectrical sampling clock pulse and ii) absorption of optical energy inthe QW channel(s) of the device from the ON pulse of the input opticalsignal produces a channel current that exceeds the bias current I_(BIAS)such that charge in the QW channel(s) of the device build to a levelthat is greater than the critical switching charge Q_(CR). When theelectrical sampling clock pulse terminates, the bias current I_(BIAS)reduces the charge in QW channel(s) of the device to a level below theholding charge Q_(H), thereby causing the device to switch from theconducting/ON state to the non-conducting/OFF state. Note that thedevice does not switch from the non-conducting/OFF state to theconducting/ON state in the event that either the electrical samplingclock pulse is not present or the input optical signal represents theOFF logic level. This occurs because these signals alone are notsufficient to produce the critical switching charge Q_(CR).

As shown in FIG. 4E, a clock generator 207′ generates an electricalclock signal that is supplied to the n-channel injector terminal(s) 308for injection into the QW channel(s) coupled thereto. The electricalclock signal includes downward running electrical clock pulses thatdefine active sampling periods whose duration overlaps the bits ofinformation encoded in the input digital optical signal as shown in FIG.4B(i). The input digital optical signals are supplied to the resonantcavities for resonant absorption by the device 201′. A plurality ofvoltage references (4 shown as 316 _(MSB), 316 _(MSB-1), 316 _(MSB-2),316 _(MSB-3)), each corresponding to a different heterojunctionthyristor device, are operably coupled to the electrical input terminal304 of the corresponding heterojunction thyristor device. The voltagereference and corresponding heterojunction thyristor-based samplingdevice cooperate to generate at the electrical output terminal 306 avoltage signal (V_(MSB), V_(MSB-1), V_(MSB-2), or V_(MSB-3))representing the contribution of the bit in the digital word inaccordance with the input digital optical signal supplied thereto.Examples of such voltage signals are shown above in table I. A summingcircuit 206′ is operably coupled to the output terminals of theheterojunction-thyristor-based sampling devices as shown. The summingcircuitry 206′ operates during the summing period to sum the voltagesignals (V_(MSB), V_(MSB-1), V_(MSB-2), and V_(MSB-3)) produced at theoutput terminals 306 of the sampling devices. The result of the summingoperation performed by the summing circuitry 206′, which is output fromthe summing circuitry 206, is an analog electric signal whose magnitudecorresponds to the digital word encoded by the bits of the input digitaloptical signals. For example, the summing circuit 206′ of FIG. 4E sumsthe four (4) output electrical signals to produce a resultant analogsignal whose magnitude corresponds to the digital word encoded by thefour (4) bits of the input digital optical signals as shown above intable II.

Preferably, the summing circuitry 206′ includes aheterojunction-thyristor-based electrically-controlled sampling deviceas described above with respect to FIGS. 4C and 4D. In thisconfiguration, the voltage signals (V_(MSB), V_(MSB-1), V_(MSB-2),V_(MSB-3)) produced at the output terminals 306 of the plurality ofheterojunction-thyristor-based sampling devices are supplied to theinput terminal 222 of the sampling device of circuit 206′, which isactivated during the summing period (which is defined by the duration ofthe summing clock pulse in the timing signal B as shown in FIG. 4B(iii))to thereby effectuate the summing operation, which produces the outputanalog electrical signal as described above. Note that theheterojunction-thyristor-based sampling device of circuit 206′ includesan input capacitance that operates to store the sum of the voltagesignals (V_(MSB), V_(MSB-1), V_(MSB-2), V_(MSB-3)) produced at theoutput terminals 306 of the plurality of heterojunction-thyristor-basedsampling devices and supplied to the input terminal 222 of the samplingdevice of circuit 206′ for output during the subsequent summing period.

Preferably, such conversion operations are repeated for another digitalword that is subsequently encoded by the input digital optical signalssupplied to the converter 201′ as described above.

In the illustrative embodiment shown in FIG. 4E, the voltage references316 _(MSB), 316 _(MSB-1, 316) _(MSB-2), 316 _(MSB-3) supply voltagelevels corresponding to maximum voltage level (V_(REF)) of the analogelectrical signal divided by 2¹, where I corresponds to the bit positionin the digital word (e.g., I=0 for MSB, I=1 for MSB-1 . . . ). However,it should be noted that alternate configurations are possible withvarying voltage reference source values.

In the embodiment of FIGS. 4A and 4E, the injector terminal of thedevice is operably coupled to the N-type QW channel(s) realized in theN-type modulation doped QW(s) structure of the device. In such aconfiguration, the sampling clock pulse of FIG. 4B(ii) is in the form ofa downward running electrical pulse supplied to the injector terminal ofthe device. In addition, the bias current source draws charge from then-type QW channel(s) to the positive supply voltage potential V_(D). Inan alternate embodiment, the injector terminal of the device may beoperably coupled to the p-type QW channel(s) realized in the P-typemodulation doped QW(s) structure of the device. In such a configuration,the sampling clock pulse of FIG. 4B(ii) is in the form of an upwardrunning electrical pulse supplied to the injector terminal of thedevice. In addition, the bias current source draws charge from thep-type QW channel(s) to ground potential. Similarly, the summing clockpulse of FIG. 4B(iii) is shown as a downward running electrical clockpulse. Such a pulse is suitable to activate a heterojunction-basedsample and hold circuit as described above with respect to FIG. 4C.Alternatively, the summing clock pulse of FIG. 4B(iii) can be an upwardrunning electrical clock pulse. Such a pulse is suitable to activate aheterojunction-thyristor-based sample and hold circuit as describedabove with respect to FIG. 4D.

Another photonic digital-to analog converter 251 is shown in FIG. 5A.Similar to the embodiment of FIG. 4A, the input digital optical signalssynchronously encode a plurality of bits of information that arelogically arranged from a most-significant-bit (MSB) to aleast-significant-bit (LSB) as shown in FIGS. 5A and 5B(i). These bitsform a digital word. In the exemplary embodiment shown, four (4) inputdigital optical signals encode four (4) bits of information logicallyarranged from a most-significant-bit (MSB) to a least-significant-bit(MSB 3). A plurality of heterojunction thyristor devices (4 shown as 252_(MSB), 252 _(MSB-1), 252 _(MSB-2), 252 _(MSB-3)), each corresponding toa different input digital optical signal/bit, are formed in resonantcavities on at least one substrate. Preferably, the plurality ofheterojunction thyristor devices are integrally formed in resonantcavities on a common substrate. In addition, a plurality of voltagedivider networks (4 shown as 254 _(MSB), 254 _(MSB-1), 254 _(MSB-2), 254_(MSB-3)), each corresponding to a different heterojunction thyristordevice and input digital optical signal/bit, are operably coupled to thecathode terminal of the heterojunction thyristor devices (252 _(MSB),252 _(MSB-1), 252 _(MSB-2), 252 _(MSB-3)). A summing circuit 256 isoperably coupled to the voltage divider networks (254 _(MSB), 254_(MSB-1), 254 _(MSB-2), 254 _(MSB-3)) as shown. An optical clockgenerator (not shown) generates a plurality of optical timing A signalsand optical timing B signals as shown in FIG. 5B(ii) and 5B(iii),respectively. Each optical timing A signal includes a sampling clockpulse whose duration defines a sampling period that overlaps the bits ofinformation encoded in the input digital optical signals. Each opticaltiming B signal includes a summing clock pulse whose duration defines asumming period that is subsequent to the sampling period.

During conversion operations, the plurality of input optical signals andthe plurality of optical timing A signals are injected into the resonantcavities, where such signals are resonantly absorbed by theheterojunction thyristor devices (252 _(MSB), 252 _(MSB-1), 252_(MSB-2), 252 _(MSB-3)). Each heterojunction thyristor device isconfigured (as described above with respect FIGS. 2B2, 2B4, 2C and 2D2)to operate during a given sampling period to convert the input digitaloptical signal supplied thereto to a corresponding digital electricalsignal. In this configuration, each heterojunction thyristor deviceswitches from the non-conducting/OFF state to the conducting/ON state inthe event that both the optical sampling clock pulse is present and theinput optical signal represents the ON logic level. This occurs becausethe combination of these signals are sufficient to produce the criticalswitching charge Q_(CR). However, each heterojunction thyristor devicedoes not switch from the non-conducting/OFF state to the conducting/ONstate in the event that either the optical sampling clock pulse is notpresent or the input optical signal represents the OFF logic level. Thisoccurs because each of these signals alone is not sufficient to producethe critical switching charge Q_(CR).

The voltage divider networks (254 _(MSB), 254 _(MSB-1), 254 _(MSB-2),254 _(MSB-3)) are configured to scale the magnitude of the digitalelectrical signals produced by the heterojunction thyristor devices (252_(MSB), 252 _(MSB-1), 252 _(MSB-2), 252 _(MSB-3)) to produce outputelectrical signals (V_(MSB), V_(MSB-1), V_(MSB-2), V_(MSB-3)) whosemagnitude corresponds to the bits encoded by the input digital opticalsignals. For example, the voltage divider networks (254 _(MSB), 254_(MSB-1), 254 _(MSB-2), 254 _(MSB-3)) shown in FIG. 5A scale themagnitude of the digital electrical signals produced by theheterojunction thyristor devices (202 _(MSB), 202 _(MSB-1), 202_(MSB-2), 202 _(MSB-3)) by factors of 1, ½, ¼, and ⅛, respectively, toproduce output electrical signals as shown in the Table I above. Thescaling function for a given voltage divider network can be representedgenerally by a scale factor of (½^(X)), where X is the differencebetween the position of the corresponding bit and the MSB.

The summing circuitry 256 operates during the summing period to sum theoutput electrical signals (V_(MSB), V_(MSB-1), V_(MSB-2), V_(MSB-3))produced by the voltage divider networks (254 _(MSB), 254 _(MSB-1), 254_(MSB-2), 254 _(MSB-3)). The result of the summing operation performedby the summing circuitry 256, which is output from the summing circuitry256, is an analog electric signal whose magnitude corresponds to thedigital word encoded by the bits of the input digital optical signals.For example, the summing circuit 256 of FIG. 5A sums the four (4) outputelectrical signals to produce a resultant analog signal whose magnitudecorresponds to the digital word encoded by the four (4) bits of theinput digital optical signals as shown in the Table II above.

Preferably, the summing circuitry 256 includes a chain of two-portadding nodes 258 and sample/hold circuits 260 arranged as pairs, eachpair correspond to a different voltage divider network (254 _(MSB-1),254 _(MSB-2), 254 _(MSB-3)) as shown. In this configuration, the outputelectrical signal generated by a given voltage divider network issupplied to an input node of the corresponding two-port adding node 258.In addition, a single sample/hold circuit 261 supplies an outputelectrical signal (V_(MSB)) whose magnitude corresponds to the MSB tothe chain of two-port adding nodes 258 and sample/hold circuits 260. Thesample and hold circuits 260 of this chain and the single sample andhold circuit 261 are activated during the summing period (which isdefined by duration of the summing clock pulse in the optical timing Bsignal as shown in FIG. 4B(iii)) to thereby effectuate the summingoperation, which produces the output analog electrical signal asdescribed above. In this configuration, each sample and hold circuit 260and 261 includes an input capacitance that operates to store theelectrical signal generated by a given voltage divider network andsupplied thereto during a given sampling period for summation during thesubsequent summing period.

Preferably, such conversion operations are repeated for another digitalword that is subsequently encoded by the input digital optical signalssupplied to the converter 251 as described above.

In another aspect of the present invention, the heterojunction thyristordevice can be configured to operate as an optically-controlled samplingdevice (e.g., optically-controlled switch), which is suitable for use ina sample and hold circuit (such as sample and hold circuits 260/261 inFIG. 5A) and in many other signal processing applications (such asswitched capacitance filters and switched capacitance waveformgenerators). Two possible configurations are shown in FIGS. 5C and 5D,respectively.

In FIG. 5C, a first p-channel injector terminal 272 (the electricalinput terminal) and a second p-channel injector terminal 274 (theelectrical output terminal) are operably coupled to opposite ends of thep-channel QW(s) of structure 20. A bias current source is coupled to then-channel injector terminal(s) 276 and draws charge from the n-type QWchannel(s) to the positive supply voltage potential V_(D). The anodeterminal 278 is forward biased (e.g. biased positively) with respect tothe cathode terminal 280. An optical clock signal is supplied to thedevice for resonant absorption in the n-type QW channel(s) of structure24 (which is part of the optical sampling region 281 as shown). Theoptical clock signal includes an optical clock pulse that defines anactive sampling period. The length and width of the device is sized suchthat it switches from a non-conducting/OFF state to a conducting/ONstate when absorption of the optical energy from the optical clock pulseproduces a channel current that exceeds the bias current I_(BIAS) suchthat charge in the n-type QW(s) of structure 24 builds to a level thatis greater than the critical switching charge Q_(CR). When the opticalclock pulse terminates, the bias current I_(BIAS) reduces the charge inthe n-type QW(s) channels to a level below the holding charge Q_(H),thereby causing the device to switch from the conducting/ON state to thenon-conducting/OFF state. Note that each heterojunction thyristor devicedoes not switch from the non-conducting/OFF state to the conducting/ONstate in the event that the optical sampling clock pulse is not present.This occurs because there is no absorption of optical energy in the QWchannel(s) of the device to produce the critical switching chargeQ_(CR).

When the device is operating in the non-conducting/OFF state, theelectrical input terminal 272 is electrically isolated from theelectrical output terminal 274. However, when the device is operating inthe conducting/ON state, the electrical input terminal 272 iselectrically coupled to the electrical output terminal 274 (and there isminimal potential voltage differences between input terminal 272 andoutput terminal 274). In this manner, the heterojunction thyristordevice operates as a sampling device (e.g., switch) that is selectivelyactivated and deactivated by an optical control signal (e.g., theoptical clock signal).

In FIG. 5D, a first n-channel injector terminal 282 (the electricalinput terminal) and a second n-channel injector terminal 284 (theelectrical output terminal) are operably coupled to opposite ends of then-channel QW(s) of structure 24. A bias current source is coupled to thep-channel injector terminal(s) 286 and draws charge from the p-type QWchannel(s) to the ground potential. The anode terminal 288 is forwardbiased (e.g. biased positively) with respect to the cathode terminal290. An optical clock signal is supplied to the device for resonantabsorption in the p-type QW channel(s) of structure 20 (which is part ofthe optical sampling region 291 as shown). The optical clock signalincludes an optical clock pulse that defines an active sampling period.The length and width of the device is sized such that it switches from anon-conducting/OFF state to a conducting/ON state when absorption of theoptical energy from the optical clock pulse produces a channel currentthat exceeds the bias current I_(BIAS) such that charge in the p-typeQW(s) of structure 20 builds to a level that is greater than thecritical switching charge Q_(CR). When the optical clock pulseterminates, the bias current I_(BIAS) reduces the charge in the p-typeQW(s) channels to a level below the holding charge Q_(H), therebycausing the device to switch from the conducting/ON state to thenon-conducting/OFF state. Note that each heterojunction thyristor devicedoes not switch from the non-conducting/OFF state to the conducting/ONstate in the event that the optical sampling clock pulse is not present.This occurs because there is no absorption of optical energy in the QWchannel(s) of the device to produce the critical switching chargeQ_(CR).

When the device is operating in the non-conducting/OFF state, theelectrical input terminal 282 is electrically isolated from theelectrical output terminal 284. However, when the device is operating inthe conducting/ON state, the electrical input terminal 282 iselectrically coupled to the electrical output terminal 284 (and there isa minimal potential voltage difference between the input terminal 282and the output terminal 284). In this manner, the heterojunctionthyristor device operates as a sampling device (e.g., switch) that isselectively activated and deactivated by an optical control signal(e.g., the optical clock signal).

Another photonic digital-to analog converter 251′ is shown in FIG. 5E.Similar to the embodiment of FIG. 5A, the input digital optical signalssynchronously encode a plurality of bits of information that arelogically arranged from a most-significant-bit (MSB) to aleast-significant-bit (LSB) as shown in FIG. 5B(i). These bits form adigital word. In the exemplary embodiment shown, four (4) input digitaloptical signals encode four (4) bits of information logically arrangedfrom a most-significant-bit (MSB) to a least-significant-bit (MSB₋₃). Aplurality of heterojunction thyristor devices (4 shown as 302 _(MSB),302 _(MSB-1), 302 _(MSB-2), 302 _(MSB-3)), each corresponding to adifferent input digital optical signal/bit, are formed in resonantcavities on at least one substrate. Preferably, the plurality ofheterojunction thyristor devices are integrally formed in resonantcavities on a common substrate. Each heterojunction thyristor device isconfigured as an optically-controlled sampling device (e.g., switch) ina manner similar to the sampling device described below with respect toFIGS. 7A and 7B, whereby the n-channel injector terminals form theelectrical input terminal 304 and the electrical output terminal 306. Abias current source is coupled to the p-channel injector terminal(s) 308and draws charge from the QW channel(s) coupled thereto. The anodeterminal 310 is forward biased (e.g. biased positively) with respect tothe cathode terminal 312. As described below in detail with FIGS. 7A and7B, the length and width of each heterojunction-thyristor-based samplingdevice are sized such that it switches from a non-conducting/OFF stateto a conducting/ON state when combination of i) absorption of opticalenergy in the QW channel(s) of the device from the sampling opticalclock pulse of the optical timing signal A and ii) absorption of opticalenergy in the QW channel(s) of the device from the ON pulse of the inputoptical signal produces a channel current that exceeds the bias currentI_(BIAS) such that charge in the QW channel(s) of the device build to alevel that is greater than the critical switching charge Q_(CR). Whenthe optical sampling clock pulse terminates, the bias current I_(BIAS)reduces the charge in QW channel(s) of the device to a level below theholding charge Q_(H), thereby causing the device to switch from theconducting/ON state to the non-conducting/OFF state. Note that thedevice does not switch from the non-conducting/OFF state to theconducting/ON state in the event that either the optical sampling clockpulse is not present or the input optical signal represents the OFFlogic level. This occurs because these signals alone are not sufficientto produce the critical switching charge Q_(CR).

As shown in FIG. 4B(ii), the optical timing signal A includes an opticalsampling clock pulse that define an active sampling period whoseduration overlaps the bits of information encoded in the input digitaloptical signal as shown in FIG. 4B(i). The optical timing signal A andthe input digital optical signals are supplied to the resonant cavitiesfor resonant absorption by the corresponding heterojunction thyristordevices as shown in FIG. 5E. A plurality of voltage references (4 shownas 316 _(MSB), 316 _(MSB-1), 316 _(MSB-2), 316 _(MSB-3)), eachcorresponding to a different heterojunction thyristor device, areoperably coupled to the electrical input terminal 304 of thecorresponding heterojunction thyristor device. The voltage reference andcorresponding heterojunction thyristor-based sampling device cooperateto generate at the electrical output terminal 306 a voltage signal(V_(MSB), V_(MSB-1), V_(MSB-2), or V_(MSB-3)) representing thecontribution of the bit in the digital word in accordance with the inputdigital optical signal supplied thereto. Examples of such voltagesignals are shown above in table I. A summing circuit 256′ is operablycoupled to the output terminals of the heterojunction-thyristor-basedsampling devices as shown. The summing circuitry 256′ operates duringthe summing period to sum the voltage signals (V_(MSB), V_(MSB-1),V_(MSB-2), and V_(MSB-3)) produced at the output terminals 306 of thesampling devices. The result of the summing operation performed by thesumming circuitry 256′, which is output from the summing circuitry 256′,is an analog electric signal whose magnitude corresponds to the digitalword encoded by the bits of the input digital optical signals. Forexample, the summing circuit 256′ of FIG. 5E sums the four (4) outputelectrical signals to produce a resultant analog signal whose magnitudecorresponds to the digital word encoded by the four (4) bits of theinput digital optical signals as shown above in table II.

Preferably, the summing circuitry 256′ includes aheterojunction-thyristor-based optically-controlled sampling device asdescribed above with respect to FIGS. 5C and 5D. In this configuration,the voltage signals (V_(MSB), V_(MSB-1), V_(MSB-2), V_(MSB-3)) producedat the output terminals 306 of the plurality ofheterojunction-thyristor-based sampling devices are supplied to theinput terminal 332 of the sampling device of circuit 256′, which isactivated during the summing period (which is defined by the duration ofthe summing clock pulse in the timing signal B as shown in FIG. 5B(iii))to thereby effectuate the summing operation, which produces the outputanalog electrical signal as described above. Note that theheterojunction-thyristor-based sampling device of circuit 256′ includesan input capacitance that operates to store the sum of the voltagesignals (V_(MSB), V_(MSB-1), V_(MSB-2), V_(MSB-3)) produced at theoutput terminals 306 of the plurality of heterojunction-thyristor-basedsampling devices and supplied to the input terminal 332 of the samplingdevice of circuit 256′ for output during the subsequent summing period.

Preferably, such conversion operations are repeated for another digitalword that is subsequently encoded by the input digital optical signalssupplied to the converter 251′ as described above.

In the illustrative embodiment shown in FIG. 5E, the voltage references316 _(MSB), 316 _(MSB-1), 316 _(MSB-2), 316 _(MSB-3) supply voltagelevels corresponding to maximum voltage level (V_(REF)) of the analogelectrical signal divided by 2^(I), where I corresponds to the bitposition in the digital word (e.g., I=0 for MSB, I=1 for MSB-1 . . . ).However, it should be noted that alternate configurations are possiblewith varying voltage reference source values.

In the embodiments of FIGS. 5A and 5E, the injector terminal of thedevice is operably coupled to N-type QW channel(s) realized in theN-type modulation doped QW(s) structure of the device. In such aconfiguration, the bias current source draws charge from the n-type QWchannel(s) to the positive supply voltage potential V_(D). In analternate embodiment, the injector terminal of the device may beoperably coupled to the p-type QW channel(s) realized in the P-typemodulation doped QW(s) structure of the device. In such a configuration,the bias current source draws charge from the p-type QW channel(s) toground potential. Similarly, the summing clock pulse of FIG. 5B(iii) isshown as a downward running electrical clock pulse. Such a pulse issuitable to activate a heterojunction-based sample and hold circuit asdescribed above with respect to FIG. 5C. Alternatively, the summingclock pulse of FIG. 4B(iii) can be an upward running electrical clockpulse. Such a pulse is suitable to activate aheterojunction-thyristor-based sample and hold circuit as describedabove with respect to FIG. 5D.

The heterojunction-thyristor-based sampling devices of FIGS. 4C-4E and5C-5E as described above may be realized with a material system based onIII-V materials (such as a GaAs/AlxGa_(1-x)As). FIGS. 3A through 3Fillustrate an exemplary structures utilizing group III-V materials forrealizing the heterojunction-thyristor-based sampling device inaccordance with the present invention. Alternatively, strained siliconheterostructures employing silicon-germanium (SiGe) layers may be usedto realize the heterojunction-thyristor-based sampling devices describedherein. Using the structure described above with respect to FIGS. 3A and3B, a heterojunction-thyristor-based sampling device can be realized asshown in FIG. 5F. This device is similar to the heterojunction thyristordevice of FIG. 3C, but includes additional mesas 294, 295 (preferablyformed by etching down near the bottom of layer 159 as shown) into whichis implanted P+ ion implants 296, 297 that form self-aligned channelcontacts to the p-type QW inversion channel(s). In this configuration,p-channel injector terminals 38C and 38D are formed via deposition ofp-type Au alloy metal layers 298, 299 on the P+ ion implants 296, 297 toform ohmic contacts thereto.

Turning now to FIGS. 6A through 7D, in another aspect of the presentinvention, a heterojunction-thyristor-based sampling device is used asthe basis for a photonic digital-to-analog converter that converts adigital word encoded by a serial digital bit stream in optical form toan output analog electrical signal corresponding to the digital word.

An exemplary embodiment of the photonic digital-to analog converter 301is shown in FIG. 6A. The serial digital bit stream is an optical signalthat sequentially encodes a plurality of bits of information which arelogically arranged from a most-significant-bit (MSB) to aleast-significant-bit (LSB) as shown in FIG. 6B(i). These bits form adigital word. In the exemplary embodiment shown, the serial digital bitstream sequentially encodes four (4) bits of information logicallyarranged from a most-significant-bit (MSB) to a least-significant-bit(MSB₋₃). A first heterojunction thyristor device 302 is formed in aresonant cavity on a substrate as shown in FIG. 6C. The firstheterojunction thyristor device is configured as an optically-controlledsampling device (e.g., optically-controlled switch) in a manner similarto the device described above with respect to FIGS. 5C, 5D, and 5E. Morespecifically, as shown in FIG. 6C, a first p-channel injector terminal304 (the electrical input terminal) and a second p-channel injectorterminal 306 (the electrical output terminal) are operably coupled toopposite ends of the p-channel QW(s) of the device 302. A bias currentsource is coupled to the n-channel injector terminal(s) 308 and drawscharge from the n-type QW channel(s) to the positive supply voltagepotential V_(D). The anode terminal 310 is forward biased (e.g. biasedpositively) with respect to the cathode terminal 312. Yet, thisconfiguration differs from that in FIGS. 5C and 5D, and 5E, in that aclock generator 314 generates an electrical clock signal that issupplied to the n-channel injector terminal(s) 308 for injection intothe n-type QW channel(s) of device 302. The electrical clock signalincludes downward running electrical clock pulses that define activesampling periods whose duration overlaps the bits of information encodedin the serial digital bit stream as shown in FIG. 6B(ii).

As shown in FIG. 6A, the serial digital bit stream is supplied to theresonant cavity of device 302 for resonant absorption by the device 302.A voltage reference 316 is operably coupled to the electrical inputterminal 304. The voltage reference 316 and the heterojunctionthyristor-based sampling device 302 cooperate to sequentially generateat the electrical output terminal 306 a voltage signal representing thecontribution of each bit of the digital word in accordance with theserial digital bit stream as described below in more detail. A summingcircuit 318, which is operably coupled to the electrical output terminal306, sequentially sums contribution of the voltage signal produced atthe output terminal 306 over the sequence of bits in the serial digitalbit stream to produce an output analog electrical signal correspondingto the digital word.

In order to perform the required sampling/switching operations, thelength and width of the device 302 is sized such that it operates duringa given sampling period defined by the electrical sampling clock asfollows. When the light level of the serial digital bit streamcorresponds to the ON logic level, channel current produced by thecombination of i) injection of electrical energy supplied by theelectrical clock pulse and ii) absorption of optical energy supplied bythe serial digital bit stream exceeds the bias current I_(BIAS) toproduce the critical switching charge Q_(CR) in the N-type modulationdoped QW structure of device 302. This causes the heterojunctionthyristor device 302 to switch to its conducting/ON state where thecurrent I through the device is substantially greater than zero butbelow the threshold for lasing I_(L). However, when the light level ofthe input digital optical signal falls to the OFF logic level, the biascurrent I_(BIAS) exceeds the channel current produced by the electricalclock pulse alone and thus draws on the injector terminal 308 to draincharge from the N-type modulation doped QW structure of device 302,which causes the channel charge to fall below the holding charge Q_(H).This causes the heterojunction thyristor device 302 to switch to itsnon-conducting/OFF state where the current I through the device issubstantially zero. When the light level of the input digital opticalsignal corresponds to the OFF logic level, the bias current I_(BIAS)exceeds the channel current produced by the combination of injection ofelectrical energy supplied by the electrical clock pulse and absorptionof optical energy supplied by the serial digital bit stream and thusdraws on the injector terminal 308 to drain charge from the N-typemodulation doped QW structure of device 302, which causes the channelcharge to remain below the holding charge Q_(H). This causes the deviceto remain in its non-conducting/OFF state where the current I throughthe device is substantially zero. When the device 302 is operating inthe non-conducting/OFF state, the electrical input terminal 304 (and thevoltage reference 316 coupled thereto) is electrically isolated from theelectrical output terminal 306. However, when the device 302 isoperating in the conducting/ON state, the electrical input terminal 304(and the voltage reference 316 coupled thereto) is electrically coupledto the electrical output terminal 306 (and there is a minimal potentialvoltage difference between the input terminal 304 and the outputterminal 306). In this manner, the device 302 operates as a samplingdevice (e.g., switch) that is selectively activated and deactivated bybinary logic level of the optical bits encoded in the serial digital bitstream, and produces at the output terminal 306 a voltage signal thatrepresenting contribution of each optical bit of the digital word.

The summing circuit 318 sequentially sums the voltage signal produced atthe output terminal 306 over the sequence of bits in the serial digitalbit stream to produce an output analog electrical signal correspondingto the digital word. Preferably, the summing circuitry 206 includes anadder node 320, a sample and hold circuit 322, and a feedback amplifier324 coupled between the sample and hold circuit 322 and the adder node320. The adder node has a first input port 326, a second input port 328,and an output port 330. The sample and hold circuit 322 has an input 332and an output 334, and is activated by the electrical clock pulse. Inthis configuration, the first input port 326 of the adder node 320 isoperably coupled to said electrical output terminal 306 of theheterojunction thyristor device 302, the output port 330 of the addernode 320 is operably coupled to the input 332 of the sample and holdcircuit 322, and the feedback amplifier 324 is operably coupled betweenthe output node 334 of the sample and hold circuit 322 and the secondinput port 328 of the adder node 320. In this configuration, at the endof the fourth (e.g., Nth) electrical clock pulse, the analog valuecorresponding to the four (N) bit digital word is produced at output 334of the sample and hold circuit 322.

In the illustrative embodiment shown in FIG. 6A, the voltage reference316 supplies a voltage level corresponding to maximum voltage level(V_(REF)) of the analog electrical signal divided by 2^((N−1)), where Nis the number of bits in the digital word. In addition, the feedbackamplifier 324 amplifies output 334 of the sample and hold circuit by afactor of 2. However, it should be noted that alternate configurationsare possible with varying voltage reference source values and feedbackamplifier gain factors. In addition, as shown in FIG. 6A. the sample andhold circuit 322 is preferably realized with an electrically-controlledheterojunction-thyristor-based sampling device as described above withrespect to FIGS. 4C and 4D.

In an alternate embodiment shown in FIG. 6D, the configuration of thedevice 302 as described above with respect to FIGS. 6A through 6C can bemodified such that the first and second n-channel injector terminals areused as electrical input and output terminals, and the electrical clocksignal injects upward running clock pulses into the p-type QW channel(s)in structure 20. In addition, the bias current draws charge from thep-type QW channel(s) to ground potential as shown. In thisconfiguration, during a given sampling period defined by the electricalsampling clock, the combination of injection of electrical energysupplied by the electrical clock pulse and absorption of optical energysupplied by the serial digital bit stream into the P-type modulationdoped QW structure 20 of the device selectively produces channel chargeabove the critical switching charge or below the holding charge suchthat device operates in the conducting/ON state and non-conducting/OFFstate, respectively.

A second exemplary embodiment of the photonic digital-to analogconverter 301 is shown in FIG. 7A. The serial digital bit stream is anoptical signal that sequentially encodes a plurality of bits ofinformation which are logically arranged from a most-significant-bit(MSB) to a least-significant-bit (LSB) as shown in FIG. 7B(i). Thesebits form a digital word. In the exemplary embodiment shown, the serialdigital bit stream sequentially encodes four (4) bits of informationlogically arranged from a most-significant-bit (MSB) to aleast-significant-bit (MSB₋₃). A first heterojunction thyristor device302 is formed in a resonant cavity on a substrate as shown in FIG. 7C.The first heterojunction thyristor device is configured as anoptically-controlled sampling device (e.g., optically-controlled switch)in a manner similar to the device described above with respect to FIGS.5C, 5D, and 5E. More specifically, as shown in FIG. 7C, a firstp-channel injector terminal 304 (the electrical input terminal) and asecond p-channel injector terminal 306 (the electrical output terminal)are operably coupled to opposite ends of the p-channel QW(s) of thedevice 302. A bias current source is coupled to the n-channel injectorterminal(s) 308 and draws charge from the n-type QW channel(s) to thepositive supply voltage potential V_(D). The anode terminal 310 isforward biased (e.g. biased positively) with respect to the cathodeterminal 312.

As shown in FIG. 7A, an optical clock signal that is supplied to thedevice for resonant absorption therein. The optical clock signalincludes optical clock pulses that define active sampling periods whoseduration overlaps the bits of information encoded in the serial digitalbit stream as shown in FIG. 7B(ii). Yet, this configuration differs fromthat in FIGS. 5C, 5D, and 5E, in that the serial digital bit stream issupplied to the resonant cavity of device 302 for resonant absorptiontherein. In addition, a voltage reference 316 is operably coupled to theelectrical input terminal 304. The voltage reference 316 and theheterojunction thyristor-based sampling device 302 cooperate tosequentially generate at the electrical output terminal 306 a voltagesignal representing the contribution of each bit of the digital word inaccordance with the serial digital bit stream as described below in moredetail. A summing circuit 318, which is operably coupled to theelectrical output terminal 306, sequentially sums the voltage signalproduced at the output terminal 306 over the sequence of bits in theserial digital bit stream to produce an output analog electrical signalcorresponding to the digital word.

In order to perform the required sampling/switching operations, thelength and width of device 302 is sized such that it operates during agiven sampling period defined by the optical clock as follows. When thelight level of the serial digital bit stream corresponds to the ON logiclevel, channel current produced by absorption of optical energy suppliedby the serial digital bit stream and the optical clock exceeds the biascurrent I_(BIAS) to produce the critical switching charge Q_(CR) in theN-type modulation doped QW structure of device 302. This causes theheterojunction thyristor device 302 to switch to its conducting/ON statewhere the current I through the device is substantially greater thanzero but below the threshold for lasing I_(L). However, when the lightlevel of the input digital optical signal falls to the OFF logic level,the bias current I_(BIAS) exceeds the channel current produced byabsorption of the optical clock alone and thus draws on the injectorterminal 308 to drain charge from the N-type modulation doped QWstructure of device 302, which causes the channel charge to fall belowthe holding charge QH. This causes the heterojunction thyristor device302 to switch to its non-conducting/OFF state where the current Ithrough the device is substantially zero. When the light level of theinput digital optical signal corresponds to the OFF logic level, thebias current I_(BIAS) exceeds the channel current produced by absorptionof optical energy supplied by the serial digital bit stream and theoptical clock and thus draws on the injector terminal 308 to draincharge from the N-type modulation doped QW structure of device 302,which causes the channel charge to remain below the holding chargeQ_(H). This causes the device to remain in its non-conducting/OFF statewhere the current I through the device is substantially zero. When thedevice 302 is operating in the non-conducting/OFF state, the electricalinput terminal 304 (and the voltage reference 316 coupled thereto) iselectrically isolated from the electrical output terminal 306. However,when the device 302 is operating in the conducting/ON state, theelectrical input terminal 304 (and the voltage reference 316 coupledthereto) is electrically coupled to the electrical output terminal 306(and there is a minimal potential voltage difference between the inputterminal 304 and the output terminal 306). In this manner, the device302 operates as a sampling device (e.g., switch) that is selectivelyactivated and deactivated by the binary logic level of the optical bitsencoded in the serial digital bit stream, and produces at the outputterminal 306 a voltage signal that represents the contribution of eachoptical bit of the digital word.

The summing circuit 318 sequentially sums the voltage signal produced atthe output terminal 306 over the sequence of bits in the serial digitalbit stream to produce an output analog electrical signal correspondingto the digital word. Preferably, the summing circuitry 206 includes anadder node 320, a sample and hold circuit 322, and a feedback amplifier324 coupled between the sample and hold circuit 322 and the adder node320. The adder node has a first input port 326, a second input port 328,and an output port 330. The sample and hold circuit 322 has an input 332and an output 334, and is activated by a clock pulse (which may beoptical or electrical in form). In this configuration, the first inputport 326 of the adder node 320 is operably coupled to the electricaloutput terminal 306 of the heterojunction thyristor device 302, theoutput port 330 of the adder node 320 is operably coupled to the input332 of the sample and hold circuit 322, and the feedback amplifier 324is operably coupled between the output node 334 of the sample and holdcircuit 322 and the second input port, 328 of the adder node 320. Inthis configuration, at the end of the fourth (e.g., Nth) clock pulse,the analog value corresponding to the four (N) bit digital word isproduced at output 334 of the sample and hold circuit 322.

In the illustrative embodiment shown in FIG. 7A, the voltage reference316 supplies a voltage level corresponding to maximum voltage level(V_(REF)) of the analog electrical signal divided by 2^((N−1)), where Nis the number of bits in the digital word. In addition, the feedbackamplifier 324 amplifies the output 334 of the sample and hold circuitryby a factor of 2. However, it should be noted that alternateconfigurations are possible with varying voltage reference source valuesand feedback amplifier gain factors. In addition, as shown in FIG. 7A,the sample and hold circuit 322 is preferably realized with anoptically-controlled heterojunction-thyristor-based sampling device asdescribed above with respect to FIGS. 5C, 5D, and 5E.

In an alternate embodiment as shown in FIG. 7D, the configuration of thedevice 302 as described above with respect to FIGS. 7A through 7C can bemodified such that the first and second n-channel injector terminals areused as electrical input and output terminals, and the bias currentdraws charge from the p-type QW channel(s) to ground potential as shown.In this configuration, during a given sampling period defined by theoptical clock, optical energy supplied by the serial digital bit streamand by the optical clock and absorbed into the P-type modulation dopedQW structure 20 of the device selectively produces channel charge abovethe critical switching charge (or below the holding charge) such thatdevice operates in the conducting/ON state (or the non-conducting/OFFstate).

The devices 302, 322, and 324 of FIGS. 6A and 7A are preferably formedon a common substrate from the same multilayer structure, such as themultilayer structures described above with respect to FIGS. 2A and 3A.In such a configuration, a plurality of transistors (such as n-typeheterojunction bipolar transistors, p-type heterojunction bipolartransistors, n-channel heterojunction FET transistors, and/or p-channelheterojunction FET transistors) may be used to build the amplifierdevice 324. A pictorial illustration of an exemplary p-typequantum-well-base bipolar transistor 802 formed from the multilayerstructure of FIG. 2A is shown in FIG. 8A. The p-type quantum-well-basebipolar transistor 802 includes a base electrode (B) electricallycoupled to the p-type QW structure 20 (preferably via spaced apartP-type implants as described above for the heterojunction thyristordevice). An emitter electrode (E) is contacted (preferably via etchingdown to the ohmic contact layer as described above for theheterojunction thyristor device) to the n-type ohmic contact layer 14. Acollector electrode (C) is electrically coupled to an n-type implant,which is electrically coupled to the n-type QW structure 24.

A plurality of such p-type quantum-well-base bipolar transistor devices802 can be configured to form a differential amplifier stage with a gainfactor of 2 as shown in FIG. 8B, which is suitable for use as theamplifier circuit 324 in the illustrative embodiment of FIG. 6A and FIG.7A. The differential amplifier stage of FIG. 8B includes anemitter-coupled pair of p-type quantum-well-base bipolar transistorswhose emitter terminals are coupled to ground through a bias currentsource as shown. The input nodes (labeled V_(A) and V_(B)) are coupledto the base electrodes of this pair of p-type quantum-well-base bipolartransistor as shown. The collector electrode of the p-typequantum-well-base bipolar transistor for the input node V_(A) is coupleddirectly to the positive power supply V_(D). The collector electrode ofthe p-type quantum-well-base bipolar transistor for the input node V_(B)is coupled to the positive power supply V_(D) through four (4)base-emitter coupled p-type quantum-well-base bipolar transistordevices. The collector terminal of the p-type quantum-well-base bipolartransistor for the input node V_(B) is coupled to the output node(labeled V_(o)). Such a configuration provides a gain factor of 2 toproduce a voltage signal at the output node of 2*(V_(A)−V_(B)).

In another aspect of the present invention, the heterojunction deviceconfigured for digital-optical-to-digital-electrical conversion asdescribed above with respect to FIGS. 2A through 2D4 can be used torealize an optical-to-digital converter as shown in FIGS. 9A through and9D(ii).

In the configuration of FIG. 9A, the heterojunction thyristor device 901is supplied a serial optical bit stream as shown in FIG. 9B(i). Duringsampling periods defined by sampling clock pulses in the electricalclock A signal as shown in FIG. 9B(ii), the heterojunction thyristordevice 902 converts the serial optical bit stream (input digital opticalsignal) supplied thereto to a corresponding digital electrical bitstream (output digital electrical signal) as described above withrespect to FIGS. 2A, 2B1, 2C, and 2D1. Optionally, a serial-to-parallelconverter 904 may be provided to convert the serial digital bit streaminto an n-bit digital word as shown.

In the configuration of FIG. 9C, the heterojunction thyristor device 901is supplied with a serial optical bit stream as shown in FIG. 9D(i).During sampling periods defined by sampling clock pulses in the opticalclock signal A as shown in FIG. 9D(ii), the heterojunction thyristordevice 902 converts the serial optical bit stream (input digital opticalsignal) supplied thereto to a corresponding digital electrical bitstream (output digital electrical signal) as described above withrespect to FIGS. 2A, 2B2, 2C and 2D2. Optionally, a serial-to-parallelconverter 904 may be provided to convert the serial digital bit streaminto an n-bit digital word as shown.

In another aspect of the present invention, the heterojunction deviceconfigured for digital-optical-to-digital-electrical conversion asdescribed above with respect to FIGS. 2A through 2D2 can be used torealize a receive module 122 in a parallel optical data link as shown inFIG. 9E. A parallel optical data link consists of a transmit module 120coupled to a receive module 122 with a multi-fiber connector 124 asshown in FIG. 1B. The transmit module typically employs an array 126 ofvertical-cavity-surface-emitting lasers (VCSELs) and a multi-channellaser driver integrated circuit 128 for driving the array of lasers toproduced a plurality of synchronous optical bit streams that aretransmitted over the multi-fiber connector 124. According to the presentinvention, the receive module 122 includes an array ofdigital-optical-to-digital-electrical converters 901 as described abovewith respect to FIGS. 9A through 9D(iii). These devices are adapted toreceive the synchronous optical bit streams and convert them intoelectrical form to produce a plurality of electrical bit streamscorresponding thereto. The plurality of electrical bit streams areprovided to one or more integrated circuits 134 that map parallel bitsencoded in the plurality of electrical bit streams into a predetermineddata format (such as a SONET frame).

Advantageously, many of the components of the optoelectronic circuitsthat perform signal conversion and signal sampling as described hereinare capable of being formed from the same multilayer structure, whichenables monolithic integration and significant cost improvements.

There have been described and illustrated herein several embodiments ofoptoelectronic circuits that perform signal conversion and signalsampling operations. While particular embodiments of the invention havebeen described, it is not intended that the invention be limitedthereto, as it is intended that the invention be as broad in scope asthe art will allow and that the specification be read likewise. Thus,while particular multi-layer structures and devices formed therefromhave been disclosed, it will be appreciated that other multi-layerstructures and devices formed therefrom can be used. Moreover, whileparticular heterojunction-thyristor configurations have been disclosed,it will be appreciated that other configurations could be used as well.It will therefore be appreciated by those skilled in the art that yetother modifications could be made to the provided invention withoutdeviating from its spirit and scope as claimed.

1. A photonic digital-to-analog converter comprising: a) a substrate; b)a resonant cavity that is formed on said substrate and into which isinjected an input optical signal, said input optical signal including aninput digital optical signal comprising a serial optical bit streamrepresenting a sequence of bits that form a digital word; and c) aheterojunction thyristor device, formed in said resonant cavity, thatincludes i) an anode terminal and a cathode terminal, ii) first andsecond channel regions disposed between said anode terminal and saidcathode terminal, and iii) an electrical input terminal and anelectrical output terminal coupled to opposite ends of said firstchannel region; d) a voltage reference, operably coupled to saidelectrical input terminal, that is adapted to operate in conjunctionwith said heterojunction thyristor device to sequentially generate atsaid electrical output terminal a voltage signal representingcontribution of each bit of said digital word in accordance with saidinput digital optical signal; and e) a summing network, operably coupledto said electrical output terminal, that sequentially sums contributionof said voltage signal over said sequence of bits to produce an outputanalog electrical signal corresponding to said digital word.
 2. Aphotonic digital-to-analog converter according to claim 1, wherein: saidsumming network includes an adding node, sample and hold circuit, and afeedback path between said sample and hold circuit and said adding node.3. A photonic digital-to-analog converter according to claim 2, wherein:said adding nodes has a first input port, second input port and outputport, said sample and hold circuit has an input and output, and whereinsaid first input port of said adding node is operably coupled to saidelectrical output terminal of said heterojunction thyristor device, saidoutput port of said adding node is operably coupled to said input ofsaid sample and hold circuit, and said feedback path is operably coupledto said second input port of said adding node.
 4. A photonicdigital-to-analog converter according to claim 3, wherein: said voltagereference supplies a voltage level corresponding to maximum voltagelevel of said analog electrical signal divided by 2(N−1), where N is thenumber of bits in said digital word, and said feedback path comprises anamplifier that amplifies output of said sample and hold circuit by afactor of
 2. 5. A photonic digital-to-analog converter according toclaim 1, wherein: in response to light intensity level of said inputoptical signal corresponding to a predetermined ON condition, charge isstored in said second channel region to cause said heterojunctionthyristor device to operate in an ON state whereby current flows betweensaid anode terminal and said cathode terminal and said electrical inputterminal is electrically coupled to said electrical output terminal. 6.A photonic digital-to-analog converter according to claim 5, wherein: inresponse to light intensity level of said input optical signalcorresponding to a predetermined OFF condition, said heterojunctionthyristor device operates in an OFF state whereby current does not flowbetween said anode terminal and said cathode terminal and saidelectrical input terminal is electrically isolated from said electricaloutput terminal.
 7. A photonic digital-to-analog converter according toclaim 6, wherein: said input optical signal further includes an opticalclock signal comprising optical clock pulses that define samplingperiods corresponding to said bits.
 8. A photonic digital-to-analogconverter according to claim 7, further comprising: a current sourceoperably coupled to said second channel region that draws charge fromsaid second channel region such that a given optical clock pulse aloneinduces a charge in said second channel region below a holding chargesuch that said heterojunction thyristor device operates in said OFFstate.
 9. A photonic digital-to-analog converter according to claim 6,wherein: an electrical clock signal is injected into said second channelregion, said electrical clock signal comprising electrical clock pulsesthat define sampling periods corresponding to said bits.
 10. A photonicdigital-to-analog converter according to claim 9, wherein: saidelectrical clock signal contributes to said charge in said secondchannel region.
 11. A photonic digital-to-analog converter according toclaim 10, further comprising: a current source operably coupled to saidsecond channel region that draws charge from said second channel regionsuch that a given electrical clock pulse alone induces a charge in saidsecond channel region below a holding charge such that saidheterojunction thyristor device operates in said OFF state.
 12. Aphotonic digital-to-analog converter according to claim 1, wherein: saidheterojunction thyristor device is formed from a multilayer structure ofgroup III-V materials.
 13. A photonic digital-to-analog converteraccording to claim 1, wherein: said heterojunction thyristor device isformed from a multilayer structure of strained silicon heterostructuresemploying silicon-germanium layers.
 14. A photonic digital-to-analogconverter according to claim 1, wherein: said heterojunction thyristordevice further comprises a p-channel FET transistor formed on saidsubstrate and an n-channel FET transistor formed atop said p-channel FETtransistor.
 15. A photonic digital-to-analog converter according toclaim 14, wherein: wherein said p-channel FET transistor comprises amodulation doped p-type quantum well structure, and wherein saidn-channel FET transistor comprises a modulation doped n-type quantumwell structure.
 16. A photonic digital-to-analog according to claim 15,wherein: said first channel region comprises at least one p-type quantumwell of said modulation doped p-type quantum well structure, and saidsecond channel region comprises at least one n-type quantum well of saidmodulation doped n-type quantum well structure.
 17. A photonicdigital-to-analog converter according to claim 15, wherein: said firstchannel region comprises at least one n-type quantum well of saidmodulation doped n-type quantum well structure, and said second channelregion comprises at least one p-type quantum well of said modulationdoped p-type quantum well structure.
 18. A photonic digital-to-analogconverter according to claim 15, wherein: said p-channel FET transistorincludes a bottom active layer operably coupled to said cathodeterminal, and said n-channel FET transistor includes a top active layeroperably coupled to said anode terminal.
 19. A photonicdigital-to-analog converter according to claim 18, wherein: saidheterojunction thyristor device further comprises an ohmic contactlayer, a metal layer for said anode terminal that is formed on saidohmic contact layer, and a plurality of p-type layers formed betweensaid ohmic contact layer and said modulation doped n-type quantum wellstructure.
 20. A photonic digital-to-analog converter according to claim1, wherein: said summer network includes another heterojunctionthyristor device performing sample and hold operations.
 21. A photonicdigital-to-analog converter according to claim 2, wherein: said sampleand hold circuit is activated by an electrical clock signal thatincludes electrical clock pulses that occur synchronously to electricalclock pulses supplied to said heterojunction thyristor device.
 22. Aphotonic digital-to-analog converter according to claim 2, wherein: saidsample and hold circuit is activated by an optical clock signal thatincludes optical clock pulses that occur synchronously to optical clockpulses supplied to said heterojunction thyristor device.